RFID Systems with Low Complexity Implementation and Pallet Coding Error Correction

ABSTRACT

Systems and methods for decoding data transmitted by RFID tags are disclosed. One embodiment of the invention includes an analyzer and equalizer configured to filter an input signal, an estimation block configured to obtain a baseband representation of the modulated data signal by mixing the filtered input signal with the carrier wave, and a coherent detector configured to perform phase and timing recovery on the modulated data signal in the presence of noise and to determine a sequence of data symbols.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/626,874, entitled “RFID Systems with Low Complexity Implementationand Pallet Coding Error Correction,” filed Feb. 19, 2015, which is acontinuation of U.S. patent application Ser. No. 14/047,989, entitled“RFID Systems with Low Complexity Implementation and Pallet Coding ErrorCorrection,” filed Oct. 7, 2013, which is a continuation of U.S. patentapplication Ser. No. 11/971,678, entitled “RFID System with LowComplexity Implementation and Pallet Coding Error Correction,” filedJan. 9, 2008, which claims the benefit of U.S. Provisional ApplicationSer. No. 60/884,197, entitled “RFID System with Low ComplexityImplementation and Pallet Coding Error Correction,” filed Jan. 9, 2007and is a continuation in part of U.S. patent application Ser. No.11/553,951, entitled “RFID Receiver,” filed Oct. 27, 2006 which claimsthe benefit of U.S. Provisional Application Ser. No. 60/731,629,entitled “RFID System,” filed Oct. 28, 2005, the disclosures of whichare incorporated herein by reference in their entireties.

BACKGROUND OF THE INVENTION 1. Field of the Invention

This invention is related to transmitter-receiver systems and inparticular is related to systems for the detection of signals indifficult environments such as for use in sensory networks and RadioFrequency Identification (RFID) systems.

2. Description of the Prior Art

The detection of signals in difficult environments, such as where thesignal to noise ratio is very low and/or the interference from othersignals is very high, has always been a substantial problem. In manysystems today classical detection theory is used in digitaltransceivers. In these systems the bit stream embedded in informationbearing signal is detected one-bit at a time using a “matched filter”designed to match the signal waveform at the input of the receiver.

What is needed is a robust and powerful method for the detection ofextremely weak signals with severe phase and timing ambiguities due tothe source characteristics and propagation environment. The proposedsystem has substantially superior performance than the classical signaldetector.

SUMMARY OF THE INVENTION

We describe methods for implementing high performance low-latency readersystems in the context of RFID inventory management systems. Principleelements of the system include a zero-intermediate frequencyarchitecture, a carrier acquisition and cancellation loop, a coarsesymbol timing recovery mechanism based on banks of parallelinterpolating and correlating filters, a coherent soft-input soft-ouputcarrier phase and timing recovery detector based on a Markov model of areceived waveform, a non-coherent soft input hard output data detector,a software programmable low-complexity transmit waveform generator, anda forward error correction encoding scheme for RFID tag data.

All blocks have been designed to provide a high level of performance (interms of the end goal of detecting signal in the presence of noise) fora given latency constraint. In this case the latency constraint isoutlined by the protocol loop imposed by RFID standards forcommunication from tag to reader to exciter to tag. Often times it isthe case that only a few tens of symbols of latency in this loop can betolerated per specifications given in related standards. One suchstandard is the EPC Global's Generation II standard (ISO Standard18000-6c) for radio frequency air interfaces.

One embodiment of the invention includes an analyzer and equalizerconfigured to filter an input signal, an estimation block configured toobtain a baseband representation of the modulated data signal by mixingthe filtered input signal with the carrier wave, and a coherent detectorconfigured to perform phase and timing recovery on the modulated datasignal in the presence of noise and to determine a sequence of datasymbols.

In a further embodiment, the analyzer and equalizer is configured tofilter at least one source of narrowband interference from the inputsignal.

In another embodiment, the analyzer and equalizer includes a low latencynotch filter, where the location of the notch can be moved to eliminatesources of narrowband interference from the input signal.

In a still further embodiment, the notch filter is implemented using afilter bank with an impulse response determined by a set of filter bankcoefficients, and the analyzer and equalizer estimates the channelimpulse response and uses it to determine the filter bank coefficients.

In still another embodiment, the notch filter is configured to adapt thelocation of the notch based upon an output of the detector.

In a yet further embodiment, the estimation block receives the carrierwave as an input.

In yet another embodiment, the estimation block is configured toestimate the frequency of the carrier wave.

In a further embodiment again, the estimation block is configured tocontrol a programmable oscillator, and the estimation block isconfigured to estimate the frequency difference between the transmittedcarrier wave and the output of the programmable oscillator and toreconfigure the programmable oscillator to reduce the frequencydifference.

In another embodiment again, the coherent detector includes a coherentdecoder that determines the sequence of symbols with the maximum aposteriori probability of having been transmitted given the data signal.

In a further additional embodiment, the coherent decoder is configuredusing a finite state machine to model the observation space.

In another additional embodiment, the finite state machine incorporatessymbol phase estimation.

In a still yet further embodiment, the finite state machine incorporatessymbol timing estimation.

In still yet another embodiment, the data signal is channel coded, andthe coherent decoder includes a soft metric estimator, a de-interleaver,a soft input soft output (SISO) decoder, an interleaver, and a channelcode decoder. In addition, the soft metric estimator is configured tocalculate initial soft metrics using the data signal and a fixed phasevalue, timing value and channel state estimated by the channel codedecoder during a previous iteration, the de-interleaver is configured tode-interleave an input generated by subtracting the output generated bythe interleaver in a previous iteration from the initial soft metrics,the SISO decoder is configured to generate updated soft metrics usingthe output of the de-interleaver, the interleaver is configured tointerleave an input generated by subtracting the output of thede-interleaver from the updated soft metrics, the channel code decoderis configured to estimate a phase value, a timing value and channelstate from the output of the interleaver, and the coherent decoder isconfigured to iterate until the initial soft metrics and the updatedsoft metrics converge.

In a still further embodiment again, the coherent decoder determines themaximum soft metric and outputs the maximum soft metric, and thecoherent decoder uses predetermined probabilities to augment at leastsome of the maximum soft metrics.

In still another embodiment again, the channel code decoder includes asoft input soft output forward error correction decoder.

In a still further additional embodiment, the sequence of symbolsincludes a preamble known by the receiver, the coherent detectorincludes an interpolator that is configured to sample and interpolatethe data signal to generate a plurality of streams possessing differentsymbol rates, the coherent detector includes a correlator that isconfigured to select a stream using at least the correlation between thestream and the known preamble, and the coherent detector is configuredto provide the selected stream to a decoder.

In still another additional embodiment, at least a portion of thesequence of symbols is constrained to a predetermined set of allowedsymbol transitions, and the correlator is configured to select a streamusing at least the correlation between the stream and the known preambleand the correlation between the stream symbol transitions and theallowed symbol transitions.

A yet further embodiment again includes an exciter in a first locationconfigured to activate an RFID tag, and a receiver in a second locationfor receiving information from an activated RFID tag. In addition, theexciter and the receiver are configured to communicate via at least onewireless link, and the receiver is configured to provide information totransmit to an activated RFID tag that is responsive to informationdecoded from signals received from the activated RFID tag to the excitervia the wireless link.

In yet another further embodiment again, the receiver is configured toperform phase and timing recovery on a signal received from an activatedRFID tag.

In a yet further additional embodiment, the receiver is configured todecode information received from an activated RFID tag by determiningthe sequence of symbols with the maximum a posteriori probability ofhaving been transmitted based upon the signal received from theactivated RFID tag.

In yet another additional embodiment, the exciter is configured toactivate an RFID tag using a signal that includes a carrier wave, andthe receiver is configured to estimate the frequency of the carrier waveand to extract the estimated carrier wave from the signal received froman activated RFID tag.

A further additional embodiment again includes a modulation encoderincluding an RF transmitter, and a digital transmit waveform generator.In addition, the digital transmit waveform generator includes a waveformlook up table that contains information concerning the shape of half ofthe waveform of a plurality of time symmetric waveforms, and themodulation encoder is configured to transmit via the RF transmitter oneof the plurality of waveforms by mirroring in time the informationconcerning the shape of one of the half waveforms contained in thewaveform look up table.

In another additional embodiment again, the digital transmit waveformgenerator further comprises a ramp-up-ramp-down block that is configuredto generate a waveform mirroring in time one of the half waveformcontained in the look up table.

An embodiment of the method of the invention includes activating theRFID tag using an exciter located in a first location, receiving amessage from the RFID tag using a receiver located in a second location,transmitting acknowledgement information from the receiver to theexciter via a wireless link, transmitting a message indicative of theacknowledgement information to the RFID tag using the exciter, andreceiving a signal from the activated RFID tag containing the dataencoded on the RFID tag using the receiver.

A further embodiment of the method of the invention also includesperforming phase and timing recovery on signals received from theactivated RFID tag by the receiver.

Another embodiment of the method of the invention also includesdetermining the sequence of symbols with the maximum a posterioriprobability of having been transmitted by the activated RFID tag basedupon the signal received from the activated RFID tag.

In a still further embodiment of the method of the invention activatingthe RFID tag further comprises activating the RFID tag using a signalincluding a carrier wave.

Still another embodiment of the method of the invention also includesestimating the carrier wave and extracting the estimated carrier wavefrom signals received from the RFID tag by the receiver and thetransmitter.

Another further embodiment of the invention includes an analyzer andequalizer configured to filter an input signal, an estimation blockconfigured to obtain the data signal by extracting the carrier wave fromthe filtered input signal, and a non-coherent detector configured toperform timing recovery on the modulated data signal in the presence ofnoise and to determine a sequence of data symbols.

In still another further embodiment, the non-coherent detector includesa non-coherent decoder that selects from the set of all possible symbolcombinations for a short sequence the symbol combination that maximizesa non-coherent combining relation.

In yet another further embodiment, the non-coherent combining relationdetermines the data symbols by selecting the values for x_(1,i) andx_(2,i) that maximize the following metric:

${Metric} = {{\prod\limits_{i = {k - N + 2}}^{k + 1}\; \left( {{r_{2,{i - 1}}x_{2,{i - 1}}} + {r_{1,i}x_{1,i}}} \right)}}$

where:

r_(1,i) is the received component in the first half of a symbol intervalafter removal of an estimate of the DC value of the received signal; and

r_(2,i) is the received component in the second half of a symbolinterval after removal of an estimate of the DC value of the receivedsignal

x_(1,i) is a hypothesis for the waveform of the first half of a symbolinterval resulting from the modulation of the data value i using themodulation scheme used to modulate the data signal onto the carrierwave; and

x_(2,i) is a hypothesis for the waveform of the second half of a symbolinterval resulting from the modulation of the data value i using themodulation scheme used to modulate the data signal onto the carrierwave.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified block diagram of an RF transmitter-receiversystem and passive sensor.

FIG. 2a is a simplified block diagram of an end to end communicationsystem of the type shown in FIG. 1.

FIG. 2b is a model of a SISO implementation of the system shown in FIG.2 a.

FIG. 3a is a diagram of a carrier offset recovery circuit to enable azero-if architecture.

FIG. 3b is a loop filter for the carrier offset recovery circuit in FIG.3 a.

FIG. 4a is a diagram of a SISO decoder as a 4-Port Device.

FIG. 4b is a block diagram of SISO processing with interleaving andde-interleaving.

FIG. 5 is an illustration of Quantized Phase Space.

FIG. 6 is an illustration of Quantized Time Domain.

FIG. 7 is an illustration of Example of Trellis Diagram.

FIG. 8a is an illustration of a Single State Trellis Transition

FIG. 8b is an illustration of a Trellis Section.

FIG. 9a is a block diagram of Single Parity Check Code (SPC).

FIG. 9b is a block diagram of an RFID SISO Decoder.

FIG. 9c represents the detailed operation of a SISO decoder forrepetition code.

FIG. 9d illustrates the operation of a SISO decoder for SPC.

FIG. 10 is a block Coherent SISO Decoder.

FIG. 11 is a block Non-Coherent SISO Decoder.

FIG. 12a illustrates a block Cascaded Non-Coherent.

FIG. 12b illustrates a Coherent SISO Decoder.

FIG. 13a is a block diagram of an RFID System.

FIG. 13b is a block diagram of a reader/interrogator of FIG. 13 a.

FIG. 14a is block diagram of FM0 encoder for RFID applications.

FIG. 14b is block diagram of Miller encoder for RFID applications.

FIG. 15a is a block diagram of classical coherent detector.

FIG. 15b is a block diagram of classical non-coherent detector.

FIG. 15c is a block diagram of a multiple symbol non-coherent detector.

FIG. 15d illustrates the operation of the multiple symbol non-coherentdetector of FIG. 15 c.

FIG. 16 shows a trellis diagram for FM0 and Miller code.

FIG. 17 shows a bit error rate as function of signal-to-noise ratio.

FIG. 18 shows a first method for a timing trellis section for pulseswith time varying duration.

FIG. 19 shows timing tick marks for the method FIG. 18.

FIG. 20 shows a second method using a folded timing trellis for pulseswith time varying duration.

FIG. 21 shows a tree diagram with three transitions per node.

FIG. 22 shows an example of a symbol tree structure method 3 for N=4,and Δmax=1.

FIG. 23 shows an example of symbol tree with windowed structure method 3for N=4, and Δmax=1.

FIG. 24 shows a SISO implementation: Intermediate metric variablecomputation.

FIG. 25 shows a SISO implementation: Interconnect of node processors andbranch select units.

FIG. 26 shows a SISO implementation: Extended parallel source nodeprocessing.

FIG. 27 shows a Forward and Backward processor.

FIG. 28 is a high-level block diagram of a symbol waveform generator.

FIG. 29 is an illustration of RFID clock and data burst recovery.

FIG. 30 is a functional architecture of the RFID clock and data burstrecovery circuit.

FIG. 31 is an illustration of the interpolator algorithm.

FIG. 32 is a detailed block diagram of the data burst recoverycorrelator.

FIG. 33 is a depiction of two applicable digital building blocks for thecorrelator.

FIG. 34 is an illustration of a pallet code technique for reading RFIDtags blocked by obstructions.

FIG. 35 is an illustration of the encoding of redundant bits for thepallet code.

FIG. 36 is a simulation of the Pallet code packet error rateperformance.

FIG. 37 is an illustration of a simple message-passing algorithm.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

Receiver subsystems may provide enhanced detection of signals where somelatency may be tolerated, particularly for use in sensory networks andpassive Radio Frequency Identification (RFID) based systems. Suchsystems may use iterative processing techniques withsoft-input-soft-output (SISO) components to combine channel decodingwith equalization, demodulation, phase tracking, symbol timing andsynchronization and interference cancellation. This is achieved withexchange of probabilities or “soft information” or equivalently theprobability of correct detection of transmitted symbols based on theobserved vector, at any given state of a finite state machine (FSM)which models the observation space. The evolution of FSM in time domainresults into a planar graph referred to here as the “Trellis”. In thepresence of additive white Gaussian noise (AWGN) with random phase andtiming, the performance of the receiver using SISO approaches that of anideal coherent receiver. In the presence of other channel anomalies suchas multipath, fading and jamming, the performance gain is much greaterthan conventional systems. The SISO decoders described here can also beused for applications where serial or parallel concatenated channelcoding methods is employed.

The system disclosed herein may use iterative algorithms that can beapplied to a broad range of sensory class of signals and waveforms.Iterative processing techniques with soft-input-soft-output (SISO)components may be used to combine channel decoding with equalization,demodulation, phase tracking, symbol timing and synchronization andinterference cancellation. This is achieved with exchange ofprobabilities or “soft information”. When the transmitted sequence isproduced from a binary symmetric source (BSS) and in presence ofadditive white Gaussian noise (AWGN), channel distortion, random phaseand synchronization error, the performance of the receiver converges tothe ideal coherent receiver for un-coded signal. In presence of otherchannel anomalies such as multipath, fading and jamming, the expectedperformance gain is much greater than conventional systems. The overallSISO decoders described here can also be used for applications whereserial or parallel concatenated channel coding methods are employed.

Referring now to FIG. 1, transmission system 1-10 transmits a signal inforward channel 1-16, such as an RF channel, which is applied to sensor1-14 which may be an RFID tag. The transmitted signal x(t) in forwardchannel 1-16 may be modeled as the real part of the complex transmittedsignal, that is x(t)=Real[a(t)e^(j(ω) ^(c) ^(t+θ))] fort∈[nT_(sym),(n+1)T_(sym)); where T_(sym) denotes the symbol timeinterval, a(t) may be complex or real-valued information bearing signaland θ denotes the phase of the transmitted signal during the symboltime. This phase can be time varying from symbol to symbol. In passiveRFID tag applications, the transmitted and received waveforms areindependent, however, the power transmitted from the tag 1-14 depends onthe power of the signal from the reader and the tag efficiency toconvert its received power to available transmit power back to thereader. In active sensors, the transmitted and received signals aretypically mutually independent signals.

Transmission system 1-10 includes data source 1-2 of transmission system1-10 is used to modulate transmitter 1-4. Antenna 1-5 applies themodulated signal through forward channel 1-16 to the sensor 1-14.Typically in RFID applications the transmitter 1-4 and the data source1-2 form the interrogator or the reader in RFID networks. Data source1-2 is used by the reader to embed an address and/or command sequence tothe device such as RFID tag 1-14. In backscatter passive RFID tags, thetransmitted signal may also embed a sinusoidal signal with a continuouswaveform (CW), which may be used to supply power to passive RFID tag.RFID tag 1-14 may then respond back with a data sequence, based on thereceived command, through the air which is illustrated as the ReturnChannel 1-18. The main function of the receiver system 1-12 is to detectthe data transmitted from the sensor 1-14, in presence of variousdistortions encountered in the return channel 1-18 such as multi-pathand/or natural and man-made interference. Receiver system includesreceiving antenna 1-7 which applies the signal received from RFID tag1-14 to receiver 1-6. The detected data from receiver 1-6 may then beconsumed by the user 1-8. In RFID applications the data user is thereader which passes the data to a higher layer protocol forinterpretation of the received packet. For passive RFID tags,transmission system 1-10 and receiver system 1-12 may be referred to as“reader/interrogator” 1-13.

The underlying transmitter receiver pair, reader/interrogator 1-13, isshown in FIG. 1. The signal is transmitted over a communication channelwith the impulse response h(t) and corrupted with additive whiteGaussian noise (AWGN) n(t), the received signal y(t) is modeled as:

y(t)=x(t)*h(t)+n(t)  (1)

where ‘*’ represents the convolution operation.

Referring now to FIG. 2, an end-to-end communication system physicalblock diagram model 2-1 for a sensory signal is shown in FIG. 2a andincludes data source 2-2 which feeds the modulator in the transmitter2-4. This signal is applied via forward channel 2-6 to sensor 2-8. Onlyin the case when the transmitted signal from sensor 2-8 is a partiallyamplified version of the original signal, the impulse responses is thecomposite impulse response of the forward and return channel, i.e.h_(f)(t)*h_(r)(t). In passive RFID applications, typically the tag mayonly use the signal from the reader to power itself. The return signalfrom the tag uses the backscatter modulation to modulate the electronicproduct code or a response back to the reader, in which case the channelimpulse response is only limited to the return channel transfer function2-10. The receiver 2-11 detects the incoming bit stream and outputs itto the user data 2-12.

In discrete time domain, we represent the sampled version of thiscomplex received signal at time n as for the k^(th) packet or frame asan N-dimensional vector,

y _(k) H _(k) x _(k) +n _(k)  (2)

Here y_(k) denotes the received complex vector with dimension N obtainedfrom uniform sampling of the received signal complex signal (afterdown-conversion) as y(nTs), where T_(s) denotes sampling interval, andthe aggregate channel transfer function represented as

$\begin{matrix}{H = \begin{pmatrix}h_{1} & 0 & L & 0 \\0 & h_{1} & h_{2} & L \\M & 0 & O & M \\0 & 0 & 0 & h_{1}\end{pmatrix}} & (3)\end{matrix}$

Channel response matrix H may be real-or-complex valued constant, orbelong to certain class of randomly distributed functions to modelin-door or out-door multi-path channel response.

The sequence error probability may be minimized, which is equivalent tomaximizing the a posteriori error probability conditioned on thesequence of observation. The estimated transmitted symbols are:

$\begin{matrix}{{\hat{a}}_{n} = {\underset{a_{n} \in \Psi}{\arg \; \max}\; {P\left( {a_{n}y} \right)}}} & (4)\end{matrix}$

where, Ψ represents the input symbol alphabet.

By applying Bayes rule we have

$\begin{matrix}{{P\left( {a_{k} = {\alpha y}} \right)} = {\sum\limits_{{\forall{a \ni a_{k}}} = \alpha}\; {\frac{P\left( {ya} \right)}{P(y)}{{P(a)}.}}}} & (5)\end{matrix}$

If Ψ={0,1} then let log likelihood ratio

$\begin{matrix}{{\Lambda_{1}\left( a_{n} \right)} = {\log {\frac{\Pr \left( {a_{n} = {1y_{k}}} \right)}{\Pr \left( {a_{n} = {0y_{k}}} \right)}.}}} & (6)\end{matrix}$

Using Bayes formula and eliminating Pr(y), we obtain reliability or“extrinsic” information

Λ₁(a _(n))=λ₁(a _(n))+λ₂(a _(n)),  (7)

where

${\lambda_{1}\left( a_{n} \right)} = {\log \frac{\Pr \left( {{ya_{n}} = 1} \right)}{\Pr \left( {{ya_{n}} = 0} \right)}}$

represents the “extrinsic information” and

${\lambda_{2}\left( a_{n} \right)} = {\log \frac{\Pr \left( {a_{n} = 1} \right)}{\Pr \left( {a_{n} = 0} \right)}}$

represents a priori log likelihood ratio (LLR) values.The sequence λ₁(a_(n)) is calculated in each iteration, and is thefunction of soft metric calculation block 4-8 shown in FIG. 4. In a SISOdecoder, such as decoder 4-2 shown in FIG. 4, the a posterioriprobability of each transmitted symbol may be computed and thensubtracted from the reliability information to remove the influence of apriori information. The extrinsic information may then be fed back (andde-interleaved if channel encoding is used) for metric calculations forthe next iteration, where:

$\begin{matrix}{{\lambda_{2}\left( a_{n} \right)} = {{{\Lambda_{2}\left( a_{n} \right)} - {\lambda_{1}^{\%}\left( a_{n} \right)}} = {\log \frac{\Pr \left( {a_{n} = {1y}} \right)}{\Pr \left( {a_{n} = {0y}} \right)}}}} & (8)\end{matrix}$

where tilde (˜) denote values from the last decoding state. In presenceof unknown random phase and timing, it may be necessary to consider theinput and output joint probability distribution functions Pr(a,φ,τ|y) inwhich the optimization problem formulated in equation (4) becomes

$\begin{matrix}{\left( {{\hat{a}}_{n},\varphi_{n},\tau_{n}} \right) = {\underset{{({a_{n},\varphi_{n},\tau_{n}})} \in \Psi}{\arg \; \max}{P\left( {a_{n},\varphi_{n},{\tau_{n}y}} \right)}}} & \left( {8a} \right)\end{matrix}$

where, Ψ represents set of all values that a_(n),ϕ_(n),τ_(n) can take.

Referring now to FIG. 2b , in theoretical model 2-30, the performance ofthe overall RFID system may be enhanced by applying a simple and novelchannel coding to the user data. This coding technique may include theuse of an outer code 2-19, the interleaver 2-18 and a single paritycheck code 2-16, defined latter which is used to drive the modulationencoder over the channel. As an example, the outer code may be arepetition code (simply taking the input data of size M and repeating itq times where q>1). The input data of size M bits may be partitionedinto N equal size subsequences each of size M/N. Each subsequence iscopied (repeated) q times (for example say q=3) then permuted bynon-identical interleavers each of size M/N, all repeated and permutedsubsequences enter a single parity check (SPC) with Nq input and oneoutput. The output sequence of SPC may be of size M/N. The call outputsequence may be considered as a parity sequence. Note that for thisexample the interleaver 2-18 is plurality of interleavers that can bemore than one. The N data subsequences and the parity sequence may thenbe multiplexed to generate a sequence of length M+M/N that enters themodulator encoder namely FM0, Miller, or any other modulator encoderused or to be used in the RFID system. It must be noted that 2-19, 2-18and 2-16 are optional in the event it is desired to attain a coding gainin RFID systems.

The SISO decoder shown may be considered to be a device that maps aninput sequence a to an output sequence c based on a finite statemachine. If the coding scheme of 2-19, 2-18 and 2-16 is employed theSISO decoder will be designed to account for the outer code 2-19,interleaver 2-18, single parity code 2-16 and modulation encoder 2-14when modeling the FSM for data encoder. The SISO decoder is embedded inthe receiver 2-28. The outer code 2-19, the interleaver 2-18 and thesingle parity code 2-16 constitute a channel coding scheme that furthertake advantage of SISO decoder for the receiver realization in 2-28. Apossible method of the channel coding technique for RFID applications isto apply the channel coding method to the RFID tag's identifier prior towriting into its memory, for passive RFID tags types that are writtenonly once and read many times. In the case of write and read many timesRFID tags, the encoder (that is 2-19, 2-18 and 2-16) can be implementedin the tag, or the reader can pre-encode the desired stored informationwhen writing into the tag. When the information is retrieved from thetag, that is when the reader reads the tag, the RFID tag transmits thestored information through channel 2-10 back to the reader. The codinggain is realized in this case by virtue of the structure of the storedinformation in the tag.

In addition, the SISO decoder may jointly estimate a random phasemodeled by ϕ in the complex multiplier in combiner 2-24 (frequency isestimated in a separated block described below). Timing offset,inherently present in any receiver subsystem and particularly inwireless systems, may be modeled in timing offset 2-26, with multi-pathpropagation characteristics. For channel model 2-22, a finite statemachine may be used to represent the channel with memory as shown inchannel matrix in equation (3).

When the data signal received by a receiver is modulated onto a carrierwave, the energy of the carrier wave is usually considerably greaterthan the energy of the data signal. Detecting the data signal can befacilitated by attempting to mix the received signal down to a zerointermediate frequency. In a number of embodiments, mixing down to azero intermediate frequency is achieved by mixing the received signalwith a signal having the same frequency as the carrier wave (i.e.extracting the carrier wave). Systems in accordance with manyembodiments of the invention utilize separate transmitters and receiversthat communicate wirelessly. As a result, the wireless receivers aretypically not synchronized to the frequency of the carrier wave of thetransmitter. A wireless receiver can attempt to estimate the frequencyof the carrier wave of a received signal and use the estimate to extractthe carrier wave from the received signal.

Referring now to FIG. 3a , a method of acquiring a carrier signal whosefrequency differs from the receiver's assumption of the transmittedcarrier by an amount defined as f_(TX)=f_(RX)=Δf is illustrated. Thissituation occurs whenever separate crystal references are used attransmitter and receiver in order to implement a target carrierfrequency (a situation that occurs when RFID exciters are not physicallywired to an RFID reader).

We note that RFID systems often operate in conjunction with a protocolthat creates a data loop from tag to reader to exciter to tag. Anothercommon constraint is that a tag will not allow more than a few tens ofsymbol times from the point of its last transmission to the point whereis receives a response from the reader through the exciter. This impliesthat latency in the aforementioned loop is an important consideration inRFID system design. Among equally performing techniques, those thatprovide lower latency are in general superior to those with higherlatency. Though it is not always explicitly mentioned, the methods andapparatus of the inventions described herein have been devised tomaximize performance for a given latency tolerance.

The received signal 3-2 (after having been mixed down by receivercarrier frequency estimate f_(RX)) is mixed again by estimateΔ{circumflex over (f)}. The result of mixing operation 3-4 is summedover N time instance 3-6, stored in a delay element 3-7 complexconjugated and multiplied 3-8 with the result of the sum over the next Ntime instances. The imaginary part of this result is taken 3-10 whichproduces error signal 3-12. The error signal is hard limited to ±1 3-14.This result is then multiplied by a micro-controlled 3-18 programmableconstant 3-16. Next, the loop filter of FIG. 3b is applied to thesignal. The loop filter has transfer characteristic

${F(z)} = {1 + \frac{b}{1 - z^{- 1}}}$

(where b is also a programmable constant). The loop filter output isthen used to control the rate at which the numerically controloscillator 3-22 oscillates to produce frequency estimate Δ{circumflexover (f)}. In other embodiments, the carrier extraction can occur usinga single mix down process with a voltage controlled oscillator that hasa center frequency tuned to f_(RX), and which is altered by the loop anamount equal to Δ{circumflex over (f)}. In many embodiments, othercircuitry is used to estimate the frequency of the carrier wave and toperform carrier wave extraction.

Referring now to FIG. 4a , a SISO decoder such as SISO decoder 4-1 canbe viewed as a four port device. The input to the SISO decoder 4-1 isthe joint probability of channel output 4-5 and transmitted symbolsequence 4-3. The output of SISO decoder 4-1 is the joint probability ofchannel output 4-7 and transmitted symbol sequence 4-11. The inputsymbol a=(a_(k)) with k∈ϰ(ϰ is the set of integers) drawn from a finitealphabet A={a₁%, a₂%, L, a_(N)%} with a-priori probability Pr(a). Letc=(c_(k)) and k∈ϰ is the sequence of output drawn from alphabet C={a₁%,a₂%, L, a_(N)%} with a priori probability Pr(c). The SISO decoder 4-1accepts at the input the sequence of probability distributions andoutputs the sequences of probability distributions, namely: the inputprobabilities P_(k)(a;I) in 4-3, P_(k)(c;I) 4-5 and output probabilitiesP_(k)(a;O)4-11, P_(k)(c;O) 4-7.

Referring now to FIG. 4b , a SISO decoder 4-2 is illustrated for usewhen the proposed channel coding scheme is employed. The input to SISOdecoder 4-2 is fed by computing equation (8), shown as the input to thede-interleaver 4-6. Input values of the De-interleaver 4-6 and theoutput of the interleaver 4-4 are the computed values from the lastdecoding iteration of the SISO decoder. The soft metric calculation maybe performed by computing equation (7) in soft metric calculator 4-8using the observed signal 4-9 and for a fixed phase value, timing andthe channel state from step 4-10. The function and structure of theinterleaver 4-4 and de-interleaver 4-6 is dictated by the repetitionrate of the outer code and discussed below. At the end of each iterationthe SISO decoder 4-2 outputs Λ₂ (a_(n)) s multiple outputs which maythen be subtracted from the output of 4-6 and fed to the interleaverblock 4-4 to compute the input metric for the next iteration, used in4-10 and 4-8. This process may be repeated until the SISO decoder 4-2converges, at which time the extrinsic information is output fordecoding the output stream.

Recursive computation of these input and output joint probabilitydistribution functions, namely: P_(k)(a;I), P_(k)(a;0), P_(k)(c;0) andP_(k)(c;I) may be made possible by modeling the received symbols asoutput from a discrete-time finite-state Markov process source. Thestate of the source at time t is denoted by S^(t) _(e), and its outputby Y. A state sequence of the source extending from time t to t′ is madepossible based on the underlying finite state machine model. Thecorresponding output forms a first order Markov chain, i.e.,

Pr(S _(θ) ^(t+1) ,S _(θ) ^(t−1) , . . . ,S _(θ) ¹)=Pr(S _(θ) ^(t+1) |S_(θ) ^(t))  (9)

Referring now to FIG. 5, for the purpose of phase sequence estimationand open loop tracking, the phase space may be quantized into Q^(ϕ)equally spaced intervals and denoted as:

$\begin{matrix}{\Theta^{\varphi} = \left\{ {0,\frac{\pi}{M},\frac{2\; \pi}{M},L,\frac{\left( {M - 1} \right)2\; \pi}{M}} \right\}} & (10)\end{matrix}$

The phase sequence can be modeled as a random walk around the unitcircle, that is a Markov process: ϕ_(n)=ϕ_(n)+Δϕ mod 2π, whereϕ_(n)∈Θ^(ϕ) and Δϕ can be modeled as discrete random variable takingvalues in the quantized phase space from a known probability densityfunction (i.e. quantized Gaussian, Tikhanov or etc.). The probability ofsuch a phase transition may be denoted as p_(ij).

The M distinct states of the Markov source are indexed by the integer m,m=0, 1, L, (M−1) with probability transition matrix:

$\begin{matrix}{P = \begin{pmatrix}p_{11} & p_{12} & L & p_{1M} \\p_{21} & p_{22} & \; & M \\M & \; & O & \; \\p_{M\; 1} & p_{M\; 2} & \; & p_{MM}\end{pmatrix}} & (11)\end{matrix}$

Where

${\sum\limits_{i = 1}^{M}\; p_{ij}} = 1$

and p_(ij)=p_(ij), i.e. matrix is symmetric and doubly Markov.

Referring now to FIG. 6, the symbol duration Q^(τ) may be quantized intoequally spaced intervals for time and synchronization. The timing spacemay be represented as:

Θ^(τ)={τ,±2τ,±3τ,L}  (12)

Let:

$\mathrm{\Upsilon} = \left\{ {\underset{i = 1}{\overset{V}{U}}\left( {t \in \left\lbrack {{{nT}_{sym} \pm {i\; \tau}},{{\left( {n + 1} \right)T_{sym}} \pm {\left( {i + 1} \right)\tau}}} \right)} \right)} \right\}$

represent the ensemble of all possible symbol timing intervals where Vis the cardinality of the set of i such that

${{{nT}_{sym} \pm {i\; \tau}}} < {\frac{\left( {{nT}_{sym} - {\left( {n + 1} \right)T_{sym}}} \right)}{2}.}$

Then J represents any member of set γ, i.e. J∈γ.

For assigning the transition probability matrix P for phase tracking, itis possible to use the classical theory of phased-lock-loops wheredistribution of state phase error and clock stability from theoscillator can be computed or estimated. Thus matrix P and can bepre-computed based on a single transition probability from one timingstate to another. For assigning the transition probability matrix P forsymbol timing, a geometric distribution can also be used (i.e.

$\frac{\partial}{M^{n}}$

where ∂ is a constant such that

$\left. {{\sum\limits_{n = 1}^{M}\; \frac{\partial}{M^{n}}} = 1} \right).$

Assuming the channel impulse response length of L, at each time instancek=1, 2, . . . , N the state of the channel is a random variable with theproperty of the memory present in the system that, given Sk, the stateSk+1 can only assume one of two values corresponding to a +1 or −1 beingfed into the tapped delay line at time k. Thus, given a binary inputalphabet {+1,−1} the channel can be in one of 2^(L) states ri, i=1, 2, .. . , 2^(L); corresponding to the 2^(L) 2 L different possible contentsof the delay elements. This set may be denoted by Θ^(h) the set ofpossible states. Additionally, let Θ^(e) represent the set of possiblestates of modulation encoder or line encoder or differential encoder.

Let the product space

Ω=Θ^(e)⊗Θ^(h)⊗Θ^(τ)⊗^(h)  (13)

represent the space of all possible states of the system, where ⊗denotes the Cartesian product and N the cardinality of Ω.

Referring now to FIG. 7, the possible evolution of states S^(n) ∈Ω canthus be described in form of a trellis diagram. An example of such atrellis structure for a 16-state trellis diagram is illustrated in FIG.7, where there are 16 transitions from each state of the trellis to theother.

The state transitions of the Markov source are governed by thetransition probabilities. In which case for the forward and backward logprobabilities of the SISO decoder may be defined as follows:

$\begin{matrix}{{{\alpha_{k}(S)} = {\underset{e:{{S^{n}{(e)}} \in \Omega}}{Max}\begin{Bmatrix}{{\alpha_{k - 1}\left( {S^{s}(e)} \right)} + {\Pi_{k}\left( {{a(e)};I} \right)} +} \\{{\Pi_{k}\left( {{c(e)};I} \right)} + {\Pi_{k}\left( {{c(e)};O} \right)}}\end{Bmatrix}}}{{\beta_{k}(S)} = {\underset{e:{{S^{n}{(e)}} \in \Omega}}{Max}\begin{Bmatrix}{{\beta_{k - 1}\left( {S^{E}(e)} \right)} + {\Pi_{k + 1}\left( {{a(e)};I} \right)} +} \\{{\Pi_{k + 1}\left( {{c(e)};I} \right)} + {\Pi_{k}\left( {{c(e)};O} \right)}}\end{Bmatrix}}}} & (14)\end{matrix}$

∀k=1, L, N, where

Π_(k)(c;I)=log(P _(k)(c;I)), Π_(k)(a;I)=log(P _(k)(a;I)

Π_(k)(c;O)=log(P _(k)(c;O)), Π_(k)(a;O)=log(P _(k)(a;O)  (15)

This maximization is over all the edges e connected to a state selectedfrom the ensemble of all possible states connected in the trellis.Equation (14) in log domain can be represented as:

$\begin{matrix}{{{\Pi_{k}\left( {c;0} \right)} = {{\underset{{e:{c{(e)}}} = c}{Max}\begin{Bmatrix}{{\alpha_{k - 1}\left( {S^{s}(e)} \right)} + {\Pi_{k}\left( {{a(e)};I} \right)} +} \\{\beta_{k}\left( {S^{E}(e)} \right)}\end{Bmatrix}} + h_{c}}}{{\Pi_{k}\left( {a;0} \right)} = {{\underset{{e:{a{(e)}}} = a}{Max}\begin{Bmatrix}{{\alpha_{k - 1}\left( {S^{s}(e)} \right)} + {\Pi_{k}\left( {{c(e)};I} \right)} +} \\{\beta_{k}\left( {S^{E}(e)} \right)}\end{Bmatrix}} + h_{u}}}} & (16)\end{matrix}$

and initial values

$\begin{matrix}{{\alpha_{0}(S)} = \left\{ {\begin{matrix}0 & {s = S_{0}} \\{- \infty} & {Otherwise}\end{matrix},{{\beta_{n}(S)} = \left\{ \begin{matrix}0 & {s = S_{n}} \\{- \infty} & {Otherwise}\end{matrix} \right.}} \right.} & (17)\end{matrix}$

The quantities h_(c) and h_(u) are normalization constants to limit therange of the numerical values of α and β. The set of states Ξ={S₁, S₂,L, S_(n)} and edges E={e₁, e₂, L, e_(k)} represent all possibletransitions between the trellis states. S^(s)(e) denotes all thestarting states for the transition e∈E to the ending state S^(E)(e) withinput symbol a(e) corresponding to the output symbol c(e).

Referring now to FIG. 8a , the operation for computation of a_(k) andβ_(k) for the binary case is illustrated in FIG. 8a , i.e. twotransitions to traverse from state S_(k) to S_(k+1). In each iteration,the forward and backward log probabilities may be computed byconsidering the Trellis structure in FIG. 8a , that is

α_(k)=max(α_(i) +m _(ik),α_(j) +m _(jk))

β_(i)=max(β_(k) +m _(ik),β_(l) +m _(il))  (18)

In order to compute the extrinsic information for each bit as shown inFIG. 8b , the input bit sequence may simply be written as:

$\begin{matrix}{{\Pi_{k}\left( {a;0} \right)} = {{\underset{\begin{matrix}{{{All}\mspace{14mu} {edges}}\rightarrow} \\{a = 1}\end{matrix}}{Max}\left\{ {\alpha + m + \beta} \right\}} - {\underset{\begin{matrix}{{{All}\mspace{14mu} {edges}}\rightarrow} \\{a = 0}\end{matrix}}{Max}\left\{ {\alpha + m + \beta} \right\}}}} & (19)\end{matrix}$

and the extrinsic for the output code is:

$\begin{matrix}{{\Pi_{k}\left( {c;0} \right)} = {{\underset{\begin{matrix}{{{All}\mspace{14mu} {edges}}\rightarrow} \\{c = 1}\end{matrix}}{Max}\left\{ {\alpha + m + \beta} \right\}} - {\underset{\begin{matrix}{{{All}\mspace{14mu} {edges}}\rightarrow} \\{c = 0}\end{matrix}}{Max}\left\{ {\alpha + m + \beta} \right\}}}} & (20)\end{matrix}$

The branch metric m is computed as:

$\begin{matrix}{m = \begin{pmatrix}{{a\; {\Pi \left( {a;I} \right)}} + {c_{1}{\Pi \left( {c_{1};I} \right)}} + L + {c_{r}{\Pi \left( {c_{r};I} \right)}} +} \\{{\Pi \left( {c_{1};\varphi} \right)} + {\Pi \left( {c_{1};\tau} \right)}}\end{pmatrix}} & (21)\end{matrix}$

where r represents the index to the selected element of channel symboloutput of c.

This establishes the equivalency of the SISO decoder as shown in FIG. 3and computation of equations (18) through (21).

Referring now to FIG. 9a , the structure of Single Parity Check (SPC)code is given. The input stream of size M is de-multiplexed in demux 9-6into blocks of N subsequences of length M/N. Each subsequencesoptionally are permuted with N interleavers each of size M/N prior toentering MUX 9-8. These “optional” interleavers (e.g., tags) togetherwith interleavers prior to single parity check code can be used toprovide security for RFID system. This method provides a highly secureRFID system The N data subsequences as described before are repeated qtimes and permuted with Nq interleavers. The output of the interleavedblocks are all exclusive OR′ed together in combiner 9-10 to form asingle parity check sequence of length M/N which is multiplexed in MUX9-2 to form a serial output stream. The outputs of multiplexer 9-2 arethen fed into the modulation encoder 9-4 (which may be FM0 or Millercode). The interleaver blocks 9-8 in this figure each are designed topreserve the Hamming weight of the input vector at the output ofinterleaver. The repetition q and the code rate defined in this case asN/(N+1) are design parameters chosen for desired length of the outputsequence and coding gain. Presently RFID tag identifiers range anywherefrom 24-bits to 2048-bits vectors, that is the output block size of theencoder. The input data U, the input to de-multiplexer 9-6, once encodedforms a vector of length M+M/N.

Referring now to FIG. 9b , the structure of Single Parity Check (SPC)decoder is illustrated as an RFID channel SIS) Decoder. The soft outputstream of size M+M/N from SISO modulation decoder is de-multiplexed indemux 9 b-6 into blocks of N+1 subsequences of length M/N. The first Nsubsequences de-interleaved through the “optional” N de-interleaverseach of size M/N which were used to provide security for RFID system.Knowing these permutation represent secure keys for RFID system, the Nsoft data subsequences after the “optional” de-interleavers may berepeated q times and permuted with Nq interleavers. The output of theinterleaved blocks are all collected together with soft outputs fromDEMUX 9 b-2 b for parity bits enter to the SISO single parity checkdecoder in SISO decoder 9 b-10 to generate soft outputs. The N+1 softoutput subsequences from SISO single parity check decoder each of lengthM/N enters Nq de-interleavers 9 b-8 b. The soft output ofde-interleavers enters the SISO repetition decoder. The output of therepetition decoder enter the “optional” interleavers. The output of the“optional” interleaver, together with soft output for parity bits fromSISO SPC, are multiplexed in MUX 9 b-2 b to form a serial output stream.The outputs of multiplexer 9 b-2 b are then fed into the SISO modulationdecoder 9 b-4 (e.g. FM0 or Miller decoder). This process goes throughseveral iterations. The output and input of SISO for repetition decoderare summed to provide reliability for input subsequences data streams.The N subsequence streams are input to de-multiplexer 9 b-6. The outputof the demultiplexed processes are input to hard decision device 9-16 togenerate decoded bit stream U. The detailed operation of SISO decoderfor repetition code is shown in FIG. 9c in steps 9 c-10, 9 c-12, 9 c-14,9 c-16 and 9 c-18. The detailed operation of SISO decoder for SPC isshown in FIG. 9d in steps 9 d-10, 9 d-12, 9 d-14 and 9 p-16.

Referring now to FIG. 10, coherent SISO receiver 10-10 is disclosed. Inthis case the extrinsic and intrinsic and branch metrics respectivelyare represented by the set of equations in equations (18) to (21).Callouts indicate the function of each processing subsystem, theanalyzer and equalizer block diagram is shown in 10-6, the estimationblock in 10-2 and the detector in 10-4. The input signal may havetraversed airspace or a wired infrastructure to reach the receiver,there is no limitation in terms of the transport mechanism into thesystem. In the particular embodiment shown, an air interface antenna10-1 passes received energy through a low-noise amplifier 10-3 and thenthrough a bandpass filter 10-5. Next, a mixer 10-7 downconverts thesignal from radio frequency to baseband to implement a zero-ifarchitecture. A key component to this is the switch DC blockingcapacitor 10-9. This capacitor, or bank of capacitors, has its switchedclosed only when a signal of interest is anticipated to be on air. Theresulting baseband signal (with DC rejected) then propagates through alow pass filter 10-11 and into a variable gain amplifier 10-13.

The received signal is then converted from an analog to a digital domain10-52. The resulting digital signal is then fed to the channel equalizerand interference canceller filter bank in input 10-8. In manyembodiments, the symbol lengths are sufficiently long that the signalexperiences very little frequency dependent fading due to multipathinterference. The signal can, however, be corrupted by narrowbandsources of interference. In one embodiment, the interference cancelingfilter consists of a low latency, yet tunable (in terms of rejectionfrequency), infinite impulse response filter that can be tuned to cancela source of narrowband interference. In many embodiments, the tunablefilter is tuned in based upon the impulse response of the channel. Inother embodiments, the tunable filter is adaptive based upon informationprovided by the SISO decoder 10-14. The output of the equalization andinterference canceller filter bank 10-8 is then rotated via a vectorphase rotation (vector complex multiplication) 10-10. The output of10-10 is used to compute the soft metric values which then feeds theSISO decoder 10-14.

The theory of operation of a coherent SISO decoder may be described asfollows: the observed vector 10-50 y is obtained from serial to parallelconversion 10-52 of the received signal to form the vector 10-50. Thesize M indicated in 10-50 is chosen as the length of the samples in thereceived packet, or for practical consideration, a convenient length forthe desired hardware complexity. The signal 10-50 is fed to channelequalizer 10-16 which is composed of a modulated filter bank. The filterbank transfer function is selected for typical deployment scenario tomatch the propagation environment and use case scenario. In particularthe filter bank can be used to excise in-band interference after theinterferer(s) has (have) been identified by the signal analysis andprediction block 10-22. The signal from channel equalizer 10-16 isrotated in phase by rotator 10-10 and fed into the correlator and softmetric estimation block 10-12. When channel coding is used, the outputof estimation block 10-12 is subtracted from the output of theinterleaver as discussed above with regard to FIG. 4 and processing inblocks 4-6 and 4-4. The de-interleaver 10-24 and interleaver block 10-26are matched to the channel encoder interleaver block used for encodingthe data in 9-8, when an optional mode when used. The signal from thede-interleaver 10-24 is fed into the SISO receiver 10-14. After eachiteration, the output of the SISO decoder 10-14 is input to theinterleaver 10-26 whose output is fed into the channel estimation block10-36 and to the metric computation block 10-12. The channel estimationblock 10-36 decodes the channel impulse response in equation (3) and isused to update the filter bank coefficients in the channel equalizer andinterference excision block 10-16. The received vector y 10-50 (alsodenoted as Y_(k) where the index k denotes the iteration index in 10-30)is also fed into clocks 10-34, 10-28, 10-39 and 10-37. In delay 10-34,the signal is delayed to match the latency required by each iteration asthe input data for the channel estimation block. In 10-37 the observedvector is used to detect the preamble sequence and initialize the symboltiming block 10-39. The symbol timing block 10-39 is updated in eachiteration from 10-62 which is the same signal as 10-60, which is theoutput of the SISO decoder. The output of the symbol timing block 10-39produces a square wave output 10-35 which is used a reference symbolclock source throughout the system. If there is a residual carrier inthe waveform, as in some RFID standards, tracking loop 10-46 is used toextract the CW and compute the phase offset from the ideal carrierfrequency in carrier offset block 10-38. If a subcarrier is used, as insome RFID standards, the output of carrier offset block 10-38 is furtherenhanced by estimating an additional phase offset term from thesubcarrier phase, by performing fine frequency tracking in 10-42 and bycomputing the phase offset in phase offset block 10-44. The frequencyestimate from fine frequency block 10-42 may a also be fed back to FFT10-18 to update the spectral estimate of the signal for the nextiteration.

In an optional mode, it may be desirable to additionally also performfrequency domain equalization in each iteration of the SISO decoder asshown in detector 10-4. This functionality may be enabled in thepresence of fast frequency fading channels in which the signal maysuffer fast fades during a single symbol interval. In an RFID system,these may be caused by conveyor belts, fast moving tunnel highways ormoving vehicles. In this case the estimated impulse response state maybe fed to the equalizer coefficient estimation block that feeds the FFTblock to compensate for fading and multipath effects.

Referring now to FIG. 11, non-coherent SISO receiver 11-10 is disclosed.The operation of all the computational blocks, namely, 11-18, 11-20,11-22, 11-16, 11-10, 11-46, 11-44, 11-42, 11-40, 11-38, 11-36, 11-34,11-37, 11-39, 11-54, 11-46, 11-40, 11-60, 11-62, are the same as in10-18, 10-20, 10-22, 10,16, 10-10, 10-46, 10-44, 10-42, 10-40, 10-38,10-36, 10-34, 10-62, 10-37 and 10-39. The channel equalizer andinterference canceller 11-16 is similar to the coherent block 10-16,except the input channel estimates may now be based on the non-coherentestimation bock for which the equations are presented below. The keydistinction between the coherent and non-coherent versions of thereceiver architecture is in the computation of extrinsic information forbranch metric computation, and the amount of phase rotation imposed inphase rotator 11-10. The channel equalizer block 11-16 is similar to thecoherent case with the exception that estimates for the channelcoefficients are derived from the non-coherent SISO detector. The SISOdecoder 11-14 uses a similar trellis to that of the coherent case,except the branch metrics computed in 11-12 are based on non-coherentsignal detection theory which is essentially phase invariant in presenceof random or unknown phase. In each iteration in block 11-12, equation(33) is computed and/or updated based on the previous iteration of theextrinsic information, that is the output from the SISO decoder's lastiteration via equation (21).

In a non-coherent case, the received signal (in absence of multipath thesymbol c_(k) is denoted simply by x_(k)) may be modeled with random orunknown phase as:

y _(k) =Ax _(k) e ^(jϕ) +n _(k)  (22)

In an AWGN channel with n_(k), that is complex zero mean Gaussian noisewith variance σ² per dimension, the observed vector's probabilitydistribution function conditioned on a known phase and the transmittedsequence of N symbols is:

$\begin{matrix}{{P\left( {{yx},\phi} \right)} = {o \cdot e^{{{- \frac{1}{\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {{y_{n} - {{Ax}_{n}e^{j\; \varphi}}}}^{2}}}\;}}} & (23)\end{matrix}$

Where o is a constant and σ is the variance of the noise. After somealgebraic manipulation we can write (23) as

$\begin{matrix}{{P\left( {{yx},\phi} \right)} = {{o \cdot e^{{- \frac{A^{2}}{2\sigma^{2}}}\; {\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}}}e^{\frac{A}{\sigma^{2}}{{Re}{({\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}e^{j\; \varphi}}})}}}}} & (24)\end{matrix}$

Averaging (24) over the uniformly distributed phase over (0,2□□ yields:

$\begin{matrix}{{P\left( {yx} \right)} = {{o^{\prime} \cdot e^{{- \frac{A^{2}}{2\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}}}{I_{0}\left( {\frac{A}{\sigma^{2}}{{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}}} \right)}}} & (25)\end{matrix}$

where I₀(⋅) represents the modified zero-th order Bessel function.

Recall

$\begin{matrix}{{P\left( {x_{i} = {xy}} \right)} = {\frac{1}{P(y)}{\sum\limits_{{x:x_{i}} = x}\; {{P\left( {yx} \right)}{\prod\limits_{l}\mspace{11mu} {P(x)}}}}}} & (26)\end{matrix}$

From (25),

$\begin{matrix}{\frac{P\left( {x_{i} = {xy}} \right)}{P\left( {x_{i} = x} \right)} = {o^{''} \cdot {\sum\limits_{{x:x_{i}} = x}{\begin{pmatrix}{e^{{- \frac{A^{2}}{2\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}} \times} \\{I_{0}\left( {\frac{A}{\sigma^{2}}{{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}}} \right)}\end{pmatrix}{\prod\limits_{l \neq i}\; {P\left( x_{l} \right)}}}}}} & (27)\end{matrix}$

Or equivalently, the extrinsic metric may be approximated in equation(27) as

$\begin{matrix}{{\prod\limits_{i}\; \left( {{x_{i} = x},0} \right)} \approx {\max\limits_{{x:x_{i}} = x}\begin{Bmatrix}{{{- \frac{A^{2}}{2\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}} +} \\{{\frac{A}{\sigma^{2}}{{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}}} +} \\{\sum\limits_{l \neq i}\; {\ln \; {p\left( x_{l} \right)}}}\end{Bmatrix}}} & (28)\end{matrix}$

For the special case when x_(n) takes values +1 and −1, then the term

${- \frac{A}{\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}$

can be ignored, since |x_(n)| is constant.

Next consider a Rayleigh fading channel model, where in equation (22),the magnitude A is Rayleigh distributed and the phase is uniformlydistributed over (0, 2π) interval. The observed vector's probabilitydistribution function conditioned on a known amplitude and thetransmitted sequence of symbols is:

$\begin{matrix}{{P\left( {{yx},A} \right)} = {{o \cdot e^{{- \frac{1}{2\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {y_{n}}^{2}}}}e^{{- \frac{A^{2}}{2\sigma^{2}}}{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}}e^{\frac{2}{2\; \sigma^{2}}{{Re}{({\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}})}}}}} & (29)\end{matrix}$

Lets assume that the average power of A is σ_(f) ² and taking theexpectation with respect to complex random variable A results inP(y|x)=E_(A) {P(y|x,A)}.

$\begin{matrix}{{P\left( {yx} \right)} = {\int{\int{{o \cdot e^{{{- {\frac{1}{2}{\lbrack{\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}}\rbrack}}}{A}^{2}} + {\frac{2}{\sigma^{2}}{{Re}{({\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}})}}}}}{dA}}}}} & (30)\end{matrix}$

which can be integrated to:

$\begin{matrix}{{P\left( {yx} \right)} = {c^{''}\frac{1}{\left\lbrack {\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}} \right\rbrack}e^{\frac{{\frac{1}{2}\lbrack{\frac{1}{\sigma^{2}}{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}}}^{2}}{\lbrack{\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}}\rbrack}}}} & (31)\end{matrix}$

For obtaining the extrinsic information, from equation (31), thefollowing equation applies

$\begin{matrix}{\frac{P\left( {x_{i} = {xy}} \right)}{P\left( {x_{i} = x} \right)} = {c^{''}{\sum\limits_{{x:x_{i}} = x}\; {\frac{1}{\left\lbrack {\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}} \right\rbrack}e^{\frac{{\frac{1}{2}\lbrack{\frac{1}{\sigma^{2}}{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}}}^{2}}{\lbrack{\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}}\rbrack}}{\prod\limits_{l \neq i}\; {P\left( x_{l} \right)}}}}}} & (32)\end{matrix}$

which after some algebraic manipulation can be simplified to:

$\begin{matrix}{{\prod\limits_{i}\; \left( {x_{i},0} \right)} = {\underset{{x:x_{i}} = x}{Max}\begin{Bmatrix}{{- {\ln \;\left\lbrack {\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}} \right\rbrack}} +} \\{{\frac{1}{2}\frac{{\frac{1}{2}\left\lbrack {\frac{1}{\sigma^{2}}{\sum\limits_{n = 1}^{N}\; {y_{n}^{*}x_{n}}}} \right.}^{2}}{\left\lbrack {\frac{\sum\limits_{n = 1}^{N}\; {x_{n}}^{2}}{\sigma^{2}} + \frac{1}{\sigma_{f}^{2}}} \right\rbrack}} +} \\{\sum\limits_{l \neq i}\; {\ln \; {P\left( x_{l} \right)}}}\end{Bmatrix}}} & (33)\end{matrix}$

It should be noted that Π_(l)(x_(l),0) is defined as ln p(x_(l))

Referring now to FIGS. 12a and 12b , and in particular FIG. 12b , theoperation of an iterative cascaded SISO receiver, with non-coherent andcoherent models as described earlier, may be as follows:

Received vector y_(k) may be initially iterated in the non-coherent SISOreceiver until the stopping rule for minimizing the sequence errorprobability is satisfied. The output of the non-coherent receiver forthe estimated symbol sequence, timing, phase and channel response maythen used for the initial estimate of these sequences in the subsequentcoherent SISO receiver. The SISO decoder of 12-14 and 12-2 areessentially same decoders, except the SISO decoder 12-12 receives itsinput branch metrics from extrinsic information from the non-coherentSISO decoder 12-6.

Coherent SISO receiver system 12-10 is shown in FIG. 12a in which areceived signal is applied to branch metric generator 12-12 whichcomputes the branch metric m_(ik) in equation (18). These branch metricsare computed for each transition for each state of the trellis fromone-to-the-other. The branch metrics are input to coherent SISO decoder12-14. After a sufficient number of iterations (indicated by loop 12-16)in effect minimizing the sequence error probability in each iteration,the decoder slightly improves and reduces the error probability. Theswitch 12-18 may driven by a fixed or dynamic rule. Typically after fiveto ten iterations, the SISO decoder output can be sampled and hardquantized. An alternative approach would be to monitor the dynamic rangeof the extrinsic values and when the values reach steady state (nolonger changing or arbitrarily small change), the SISO decoder may bestopped and the output may be hard limited in block 12-20 and decoded.

Referring now to FIG. 12b , cascaded non-coherent and coherent SISOreceiver system 12-22 is depicted in which the received signal is usedby the branch metric generator 12-8 to output the metric values to thenon-coherent SISO decoder 12-6. The difference between 12-2 and 12-8 isthat the branch metric values for the non-coherent case are computed byconsidering the distribution of phase to be random, where in thecoherent case the phase is assumed to be known. After a sufficientnumber of iterations (as shown by loop 12-24), the output of thenon-coherent SISO decoder 12-6 is sampled by sample switch 12-26 andapplied to coherent SISO decoder 12-12. After a sufficient number ofiterations (as shown by sample loop 12-28), the output is applied bysample switch 12-30 to hard limiter 12-4 and then provided to the datauser which in RFID application is the protocol layer-2 embedded in thereader system. An option for the stopping rule to close the switches12-26, 12-28 and 12-30 is to monitor the rate of growth of theaccumulated forward and backward metrics in equation (14) in each SISOdecoder, and then stop the iteration when the difference betweensuccessive iterations is arbitrarily small for all the states in theTrellis.

Referring now to FIG. 13a , an implementation of RFID system 13-10 isshown in which plurality of inventory items, such as items 13-7, each ofwhich may include a passive RFID tags 13-6. The reader/interrogator 13-1emanates a signal to RFID tags 13-6 to respond with their respectiveidentification code referred to as “electronic product codes” (EPC). TheRFID tags 13-6 subsequently respond to signals generated by reader 13-1by backscattering the received signal with their respective EPCs. Thesignal may be corrupted by multipath 13-4 from the flooring the wallsand moving and fixed obstacles 13-8.

Referring now to FIG. 13b , reader/interrogator 13-1 includestransmitter-antenna subsystem 13-24 which is modulated by data from datasource system 13-36 applied to modulation encoder 13-15 and transmitsencoded RF signals to RFID tags 13-7. Alternatively, outer coder 13-28,interleaver 13-48 and/or single parity coder 13-50 may be insertedbetween data source 13-36 and encoder 13-52 in appropriateimplementations.

Reader interrogator 13-1 also includes receiver 13-28 which receivesreflected signals from RFID tags 7 and applies them to receiver system13-30 for detection and decoding. The output of receiver system 13-30provides data to a system user. Various obstacles, including sources ofinterference and multi-path reflection such as stationary and movingobjects 13-8 may be in the path of the transmitted and/or receivedsignals.

System 13-10 may, for example, be deployed in a department store inwhich items in cases of items with RFID tags 13-6 are displayed forsale. Moving and stationary objects 13-8 may represent shoppers andstore personnel moving during the day across the transmission andreception paths as well as relatively stationary objects such as one ormore potential reflectors, e.g., including advertising displays or otherracks of may be moved or changed on a less frequent basis. System 13-10may be deployed in order to keep track of inventory items, for example,to detect and prevent attempted shoplifting and/or for other reasonsrelated to inventory control by providing data from reader interrogator13-1 to a user.

Data source system 13-12 includes data source 13-36 which provides acommand or an EPC to the RFID tag 13-7. Channel coding techniques may beused to improve the overall system performance. In the event channelcoding is employed, the electronic product code stored in RFID tag 7 isused as the input. The data source may embed the Electronic Product Code(EPC) in pre-coded format by using outer coder 13-38, interleaver 13-48and single parity coder 13-50 at the time that the stored data iswritten into the RFID tag. In that event, the model of data source 13-36simplifies to a table look-up for EPC which, in the case of passive RFIDtags, is backscattered (or transmitted) to the interrogator or the RFIDreader. It is also noted that it is possible to compress the EPC code inthe interrogator prior to application of channel coding, using simplehashing method to decrease the length of the information sequence. Thus,the additional storage in the RFID tag is used to store the resultingparity bits. For example, for a 32-bit EPC code, the sequence can behashed into a 16-bit code by the interrogator in which case theeffective coding rate would be rate one-half, and the remaining 16-bitis used as a parity check sequence. The channel coding method may beused for protection against channel error. The code shown in FIG. 9a isa repetition code in which each data bit is repeated with a fixed numberof multiplicity. The number of repetitions of each input bit is a designparameter which determines the overall coding rate of the system. Theinterleaver 13-48 serves to permutate the encoded data from the outercode output 13-38, such that the Hamming distance of the input sequenceis preserved and the output is applied to single parity coder 13-50 anddescribed earlier in FIG. 9a . The output of which is applied tomodulation encoder 13-52. The modulator may use various modulationtechniques and waveforms defined by various RFID tag or sensorystandardization bodies. System 13-10 is applicable to any modulationtechnique or waveform. That is, modulation encoder 13-52 can beamplitude shift keying (ASK), on-off keying (OOK), frequency modulation(FM) and other modulation schemes without any loss of generality inapplying the receiver subsystem in 13-30.

RFID tags 13-7 operate to backscatter or actively transmit the embeddedinformation sequence to the reader or the interrogator to produce thesignals received in equation (2). Stationary and moving obstacles 13-8may cause interference between the signals reflected by one or more ofthe RFID tags 13-7. The signals received by receiver 13-30 may thereforeinclude channel interference, multi-path reflections and other effectswhich make detection and discrimination between RFID tags 13-7 difficultwith very low signal levels. Additional anomalies may also be present inthe communication channel. In an in-door environment such as warehouses,factories and malls substantial scattering due multipath and man-madeinterference (e.g. drill noise, cordless phones) or naturalinterferences (e.g. ceiling lighting) may also be present. In an outdoorenvironment interference and multipath effects may also be present inaddition to signal blockage due to foliage and weather effects due tohumidity or rain. These channel anomalies and interferences may all behandled by receiver system 13-30.

Receiver system 13-30 serves to detect and discriminate between RFIDtags 13-7 by taking advantage of SISO decoding proposed herein. It isassumed that latency in SISO decoding can be tolerated by the users ofthe system 13-10 or the processing time in system 13-10 is short forreal or near real time detection of motion of individual RFID 13-7. Thatis, reader/interrogator 13-1 processes the signals received by receiversystem 13-30 for a relatively long time and is able to distinguishbetween transmission channels from different RFID tags by learning thesignal characteristics. Furthermore, in warehouse, factory and airportdeployment scenarios the RFID tags may be moving at high velocity onconveyor belts or moving vehicles while being manipulated. ReaderInterrogator 13-1 also compensates for effect of moving RFID tags on thecharacteristic of the received signal. That is the system 13-10 cantolerate a high level of Doppler shift and still achieve highperformance in detection of signals from RFID tags.

Receiver system 13-30 also provides a frequency signal to receiver 13-28to adjust the transmitted frequency. This frequency adjustment providesa mechanism to both track and adopt the frequency channel used to readand write information into the RFID tag and also support optionalwaveforms which employ frequency hopping techniques as defined by RFIDStandardization Bodies (e.g. ISO, EPC Global).

Receiver system 13-30 processes the received signals in channelequalizer and interference canceller 13-54 which is realized by a bankof adaptive linear phase filter banks with the objective of maximizingthe received signal power and minimizing the effect of interference byexcision. That is, the frequency response of the filter bank is designedto eliminate narrow band interference while maximizing thesignal-to-noise ratios received from the RFID tag. The output of channelequalizer and interference canceller 13-56 is applied to rotator 13-58which appropriately adjust the phase of the incoming signal in real-timeto track the phase of the incoming signal and compensate for the effectof motion, Doppler and phase noise due to imperfection in theenvironment.

The output of rotator 13-56 is applied to SISO processor 13-58, whichincludes SISO decoder 13-60, and which serves to input into the softmetric calculator which calculates the intrinsic metric valuesassociated with each transition from one state of trellis structure tothe other state (or Bi-partite graph). The output of SISO processor13-58 is applied to phase, channel and frequency estimator 13-62 whichserves to provide an instantaneous phase and frequency estimate of thereceived signal based on the output of the SISO decoder.

One output of phase, channel and frequency estimator 13-62 is applied torotator 13-56 and provides the reference for phase compensation of thereceived signal which may have been caused by motion or other anomalies.A second output of phase, channel and frequency estimator 13-62 isapplied to channel equalizer and interference canceller 13-54 andprovides the adaptation algorithm with the phase and frequency ofvariables used to compute the channel equalizer and interferencecanceller coefficients of the Finite Impulse Response Filter.

The output of user system 13-34 is formed by summing the extrinsic andintrinsic information and then hard quantizing the resulting sum. Thisvalue constitutes the detected bit stream from the RFID tag. SISOdecoder 13-60 in SISO processor 13-58 also includes soft metricscalculator 13-66, de-interleaver 13-64 and interleaver 13-67 whenchannel coding as discussed earlier is employed.

In operation of system 13-30, both fine and coarse motion of items 13-6having RFID tags 13-6 can be accurately detected and observed becausethe SISO decoder in 13-60 simultaneously estimates the channel response,symbol timing and effectively performs open loop phase tracking.

Referring now to FIGS. 14a and 14b , the FM0 and Miller codes can alsobe used in passive RFID applications. In FM0 encoder 1410, user data14-22 is provided to x-or gate 14-18 the output of which is provided tobit mapper 14-16 as well as to simple delay circuit 14-23. The output ofdelay 14-23 may be provided as a second input to x-or gate 14-18 as wellas to inverter 14-20, the output of which is applied to bit mapper14-14. The two sequences from bit mappers 14-14 and 14-16 arerespectively multiplexed into a single stream in multiplexer 14-12 andeach repeated once in repeater 14-21 to provide output data 14-24 whichis the FM0 encoded data for use by the modulator. FM0 encoder 1410 maybe used as a data source, such as data source 2-2 in FIG. 2 or datasource 2-20 in FIG. 2 b.

In Miller encoder 14-60, delay taps 14-44 and 14-36 are used. Thecombinatorial logic consists of three inverters 14-46, 14-50, 14-52, twoor-gates 14-32 and 14-34, and three and-gates 14-42, 14-36 and 14-38.The output sequence of the combinatorial and delay logic is fed into thesignal mapping blocks 14-56 and 14-58 which is multiplexed in 14-30 andrepeated in 14-57 to form the output data 14-48 for Miller encoded datafor use by the modulators.

Referring now to FIG. 15a , the technique for the coherent detection ofFM0 or Miller encoded signal is depicted. The system consists of usingthe channel data 15-28 and integrating the signal over each symbolperiod during each half a symbol interval in integrate and dump 15-34and de-multiplexing the real and imaginary part in demux 15-36 andforming the cross product in step 15-38, taking the real part in block15-42, hard quantizing the output in quantizer 15-44 and mapping thedata into ones and zeros from −1/+1 in data block 15-46. This is thecoherent case.

Referring now to FIG. 15b , a similar operation is performed innon-coherent case in blocks 15-30, 15-32, 15-60, 15-62, 15-64 and 15-52except only the real part of the signal is used to form the crossproduct. The output of the non-coherent detector 15-66 and coherentdetector is 15-48. N_(s) denotes the number of sample per symbol, i.e.N_(s)=T_(sym)/T_(s) that is the ratio of the symbol time to samplingperiod.

Referring now to FIG. 15c , a block diagram of a multiple symboldetector is shown for the non-coherent detection case. Channel data15-80 is applied to integrate and dump 15-80 and the real part isdetermined by block 15-76 and demultiplexed in demux 15-74. The outputis applied to non-coherent multiple symbol detector 15-70 to provideoutput 15-72.

Referring now to FIG. 16, multiple symbol non-coherent detector(MSNNonCoh) for FM0 is disclosed. In this example, the non-coherentdetection is done over a particular sequence of half symbol observationswhich starts from middle of data interval rather than beginning of datainterval. In passive RFID systems, the presence of CW translates intohaving a DC component. The received signal with FM0 encoding at time iin presence of DC component is:

y _(1,i)=(x _(1,i) +c)e ^(jϕ) +n _(1,i)

i=k−N+2,L,k+1  (33)

y _(2,i)=(x _(2,i) +c)e ^(jϕ) +n _(2,j)

where c is a dc component, ϕ is carrier phase offset (or phase noise).Without loss of generality, it can be assumed that the phase error isconstant over duration of a symbol and n_(1,i) and n_(2,i), zero meancomplex Gaussian samples with variance □² per dimension. First the dcvalue c at the receiver can be estimated as:

$\begin{matrix}{\hat{c} = {\frac{1}{N}\left( {\sum\limits_{i = {k - N + 2}}^{k + 1}\; \left( {y_{2,{i - 1}} + y_{1,i}} \right)} \right)}} & (34)\end{matrix}$

The new observation can be defined as

r _(1,i) =y _(1,i) −ĉ

r _(2,i) =y _(2,i) −ĉ  (35)

Next the Maximum Likelihood (ML) probability of the modified observationcan be computed. The particular observations are:

r _(2,k−N+1) ,r _(1,k−N+2) ,r _(2,k−N+2) ,r _(1,k−N+3) , . . . ,r _(2,k),r _(1,k+1)

The conditional probability function can be formulated as:

$\begin{matrix}{{P\left( {{rx},\varphi} \right)} = {{constant}\; e^{{- \frac{1}{2\; o^{2}}}{\sum\limits_{i = {k - N + 2}}^{k + 1}\; {\{{{{r_{2,{i - 1}} - {x_{2,{i - 1}}e^{j\; \varphi}}}}^{2} + {{r_{1,i} - {x_{1,i}e^{j\; \varphi}}}}^{2}}\}}}}}} & (36)\end{matrix}$

Averaging (36) over the carrier phase, the ML function can beapproximated with

$\begin{matrix}{{P\left( {rx} \right)} \approx {I_{o}\left( {\frac{1}{2\; \sigma^{2}}{{\sum\limits_{i = {k - N + 2}}^{k + 1}\; \left\{ {{r_{2,{i - 1}}x_{2,{i - 1}}} + {r_{1,i}x_{1,i}}} \right\}}}} \right)}} & (37)\end{matrix}$

Since the zero order modified Bessel function is a monotonic function,thus the required metric for the decision is:

$\begin{matrix}{{Metric} = {{\sum\limits_{i = {k - N + 2}}^{k + 1}\left( {{r_{2,{i - 1}}x_{2,{i - 1}}} + {r_{1,i}x_{1,i}}} \right)}}} & (38)\end{matrix}$

The key property of FM0 encoder output 14-24 for RFID application is

x _(2,i) =d _(i) x _(2,i−1)

x _(1,i) =−x _(2,i−1)  (39)

Replacing (39) in (38) one gets:

$\begin{matrix}{\mspace{79mu} {{Metric} = {{{\sum\limits_{i = {k - N + 2}}^{k + 1}{\left( {r_{2,{i - 1}} - r_{1,i}} \right)x_{2,i,{- 1}}}}}\mspace{14mu} {or}}}} & (40) \\{\left( {{\hat{d}}_{k},{\hat{d}}_{k - 1},\ldots \mspace{14mu},{\hat{d}}_{k - N}} \right) = {\arg \; {\max\limits_{d_{k},d_{k - 1},\ldots \mspace{14mu},d_{k - N + 2}}{{\left( {r_{2,{k - N + 1}} - r_{1,{k - N + 2}}} \right) + {\sum\limits_{i = {k - N + 3}}^{k + 1}\; {\left( {r_{2,{i - 1}} - r_{1,i}} \right){\prod\limits_{j = {k - N + 2}}^{i - 1}\; d_{j}}}}}}}}} & \left( {40a} \right)\end{matrix}$

By expanding the sum for the case of N=3, the optimum decision rule formultiple symbol non coherent detection rule becomes:

$\begin{matrix}{\left( {d_{k},d_{k - 1}} \right) = {\arg \; {\max\limits_{d_{k},{d_{k} - 1}}{{{\left( {r_{2,{k - 2}} - r_{1,{k - 1}}} \right)d_{k - 1}} + \left( {r_{2,{k - 1}} - r_{1,k}} \right) + {\left( {r_{2,k} - r_{1,{k + 1}}} \right)d_{k}}}}}}} & (41)\end{matrix}$

In step 15-70, equation (41) is realized and the maximizationimplemented over all possible data symbols d_(k) ∈{+1,−1}. Note thatthis metric is independent of dc offset and can be used even withoutusing (39).

Referring now to FIG. 16, the Trellis diagrams from the encoderstructure of FIG. 14 is shown in trellis 16-2 for FM0 and in trellis16-4 for Miller code. These Trellis diagrams are used for the SISOdecoder for application of passive RFID tag standards which employ theseencoding techniques. In the presence of random phase and timing, theTrellis diagram may be too large and impractical to be illustratedgraphically. That is due to the large number of states and transitions(e.g. 2000 states and 20 transitions per state) which depends on thechoice of the cardinality of the sets in equations (10) and (12).

Referring now to FIG. 17, the performance of detectors in FIGS. 15a, 15band 15c are compared. The theoretical performance of coherent andnon-coherent detector over AWGN are also depicted with solid lines andthe performance of multiple symbol non-coherent MSNC is simulated andshown by small triangles. It is noted that the performance of the symbolnon-coherent detector outperforms the classical non-coherent detector bya factor of 3.5 dB.

In SISO decoder 4-2 a long sequence, typically a packet or a frame, isprocessed at a time. Hence, the performance is still even superior tothat of symbol non-coherent detection in which the Maximum Likelihooddetector is considering only three symbols. In applications for system13-10 are for RFID tag standards in which the tag protocol is amenableto longer latency which results from the SISO decoding. Typicalapplications of multiple symbol non-coherent detector 15-70 are when theRFID standard requires very strict timing requirements between thetag-to-reader and reader-to-tag packet inter-arrival time. These tighttimings requirements typically occur when acknowledgement or replies arerequired from the reader to the RFID tag or vice versa. In certaincircumstances and for certain RFID tag standards, it is also possible toemploy both systems 13-10 and 15-70 so that for certain packets typeswhich the system 15-70 is used and system 13-10 may be used for othertiming critical packets. Specifically, when detecting the product codeitself from the received packet, SISO decoding in 13-10 may be used, butfor other packet types which are short replies and handshake, system15-70 may be used.

Referring now to FIGS. 18 and 19, a typical problem for many digitalcommunication systems is that the baud rate is fixed, i.e. thetransmitted pulse duration is fixed. In sensor networks and RFID systemsin particular, the transmitted pulses from the tag can change induration from symbol to symbol. Thus at the reader's receiver, theduration of pulses or the instantaneous time varying baud rate should betracked. In order to track timing for such applications, a timingtrellis may be used.

Assume that the time axis is sampled at time instants iTs, where 1/Tsrepresent the sampling rate. The sampling points at the output ofmatched filter are depicted by bullets “•” on the time axis in FIG.19-10. There may be N samples per nominal baud interval. Thus thenominal symbol duration would be T₊=NTs, where T is the nominal symbolduration. Assume the symbol duration from symbol to symbol can increase(as a result for example of changing channel characteristics) by onesample to T⁻=(N+1)Ts with probability p, or no change T=NTs withprobability (1−2p), or can decrease by one sample to T=(N−1)Ts withprobability p.

Each interval has N tick marks that denote possible positions (inmultiples of Ts), where a sample can be taken at the output of thematched filter. From FIG. 19, some intervals have one tick mark asdepicted in 19-2, 19-3 and 19-4, some intervals have two tick marks,while some intervals have no tick marks as shown in 19-12. Denote thetiming state as Sk which its value corresponds to a tick mark. The stateis associated with a time interval ((k−1)T, kT], and can take one of thevalues in the following set: Sk=s0, s1, . . . , sN, sN+1. State s0denotes that the kth symbol interval ((k−1)T, kT] is not sampled at all.For example, in FIG. 19, the interval corresponding to Sk=s2 shown in19-13 is sampled at the second tick from the start of the interval StateSk=si, for 1≤i≤N denotes that the kth symbol interval ((k−1)T, kT issampled only once at the i-th tick.

Referring now also to FIG. 18, state Sk=sN+1 (for N=8 in FIG. 19)denotes that the kth symbol interval ((k−1)T, kT] is sampled twice. Theonly way an interval can be sampled twice is if it is sampled at thefirst and Nth ticks, since we only allowed maximum of one samplevariation from symbol to symbol. The constraints prevent any other wayof two samples falling in the same interval. There are some restrictionson how the sampling-states Sk can evolve. To represent all validsampling-state transitions, we form a timing trellis, depicted in FIG.18. To a branch in the timing trellis, we associate a transitionprobability Pr(Sk|Sk−1). The transition probabilities can be computedbased on parameter p. A key feature of the timing trellis in FIG. 18 isthat the branches in the trellis carry a variable number of samples. Wewill denote the vector of samples taken in the timing interval ((k−1)T,kT] by r k. Note that r k could be an empty vector if no sample is takenin the kth symbol interval.

The timing trellis has N+2 states. The present state S_(k−1) i.e. Sothrough S₉ (18-12) and next states S_(k) i.e. states S₀ (18-14) through(18-16) are shown in FIG. 18. Lets assume N=8 for clarification andrectangular NRZ pulses. For symbols with nominal duration of 8 samples,the matched filter sums the recent 8 samples. To transition from stateS₀ to state 1, the matched filter sums nine recent samples and producesobservation rk. To transition from state S₀ to state 9, the matchedfilter sums nine recent samples and produces observation r_(k), and thenext seven samples to produce observation rk+1 corresponding to data akand ak+1 respectively. Each state also should store the most recentindex of observation sample. The branch metric per edge of trellisrequires variable number of samples. Denote the index of observationsample to state S_(k−1) at time k−1 by q_(i), then the number of samplesrequired to compute the edge branch metric are q_(i)+N(s_(i),s_(j))samples. Where N(s_(i),s_(j)) represent the number of samples requiredto compute the branch metric from present state S_(k−1)=s_(i) to nextstate S_(k−1)=s_(j), For N=8:

N(s0,s1)=9 N(s0,s9)=9+7 N(s1,s1)=8 N(s1,s2)=9 N(s1,s9)=8+7

N(s2,s1)=7 N(s2,s2)=8 N(s2,s3)=9 N(s2,s9)=7+7 N(s2+i,s2+i−1)=7 for i=1,2, 3, 4, 5N(s2+i,s2+i)=8 for i=1, 2, 3, 4, 5 N(s2+i,s2+i+1)=9 for i=1, 2, 3, 4, 5

-   -   N(s8,s0)=no samples        N(s8,s7)=7 N(s8,s8)=8 N(s9,s0)=no samples N(s9,s7)=7 N(s9,s8)=8

Referring now to FIG. 20, a folded timing trellis is similar to themethod shown in FIGS. 18 and 19, except the number states are less andnumber of transitions per state is fixed number for all timing states.The time axis may be sampled at time instants iTs, where 1/Ts representthe sampling rate. The sampling points at the output of matched filter(end of actual symbol time duration) are depicted by bullets “•” on thetime axis in FIG. 19-10. Assume there are N samples per nominal baudinterval. Thus the nominal symbol duration is T₊=NTs, where T is thenominal symbol duration. Assume the symbol duration from symbol tosymbol can increase by one sample to T−=(N+1)Ts with probability p, orno change T=NTs with probability (1−2p), or can decrease by one sampleto T=(N−1)Ts with probability p.

Each interval has N tick marks that denote possible positions (inmultiples of Ts), where a sample can be taken at the output of thematched filter. The state is associated with a time interval ((k−1)T,kT], and can take one of the values in the following set: Sk=s1, . . . ,sN. State Sk=si, for 1≤i≤N denotes that the kth symbol interval ((k−1)T,kT] is sampled at the i-th tick from the beginning of the interval.There are some restrictions on how the sampling-states Sk can evolve. Torepresent all valid sampling-state transitions, a timing trellis may beformed as depicted in FIG. 20. To a branch in the timing trellis, aprobability Pr(Sk|Sk−1) can be assigned to a transition. The transitionprobabilities can be computed based on parameter p. A key feature of thetiming trellis in FIG. 20 is that the branches in the trellis carry avariable number of samples. The vector of samples taken in the timinginterval ((k−1)T, kT] can be denoted by r k.

The timing trellis has N states. Lets assume N=8 for clarification. Thepresent states S_(k−1) i.e. S₁ (20-10) through S₈ (20-12) and nextstates S_(k) i.e. states S₁ (20-14) through (20-16) are shown in FIG.20. Also assume rectangular pulses. For symbols with nominal duration of8 samples, the matched filter sums the recent 8 samples if there is atransition from present state Si to the next state Si for i=1, 2, . . ., N (for the example in FIG. 20, N=8). For symbols with nominal durationof 8 samples, the matched filter sums the recent 9 samples if there is atransition from present state Si to the next state Si+1 for i=2, 3, . .. , N, and S₁ to S₈ (for the example in FIG. 20, N=8). For symbols withnominal duration of 8 samples, the matched filter sums the recent 7samples if there is a transition from present state Si to the next stateSi−1 for i=1, 2, . . . , N−1, and from present state SN to S1 (for theexample in FIG. 20, N=8). Each state also should store the most recentindex of observation sample to compute the branch metric for the nexttrellis section. Thus the branch metric per edge of trellis requiresvariable number of samples. Denote the index of observation sample tostate Sk−1 at time k−1 by q_(i), then the number of samples required tocompute the edge branch metric are q_(i)+N(s_(i),s_(j)) samples. WhereN(s_(i),s_(j)) represent the number of samples required to compute thebranch metric from present state S_(k−1)=s_(i) to next stateS_(k−1)=s_(j), For N=8:

N(s1,s2)=7 N(s1,s1)=8 N(s1,s8)=9

N(si,si−1)=7 for i=1, 2, 3, 4, 5, 6, 7N(si,si)=8 for i=1, 2, 3, 4, 5, 6, 7, 8N(si,si+1)=9 for i=2, 3, 4, 5, 6, 7, 8

N(s8,s1)=7 N(s8,s8)=8 N(s8,s7)=9

Referring now to FIGS. 21 and 22, the symbol timing tree structure isbased on a nominal value of N samples per symbol. This means that thegranularity of the timing captured will be 1/N×F_(s) where F_(s) is thesampling frequency. The rates may be discussed in the 1/F_(s) domain,meaning that rates will be expressed in percentage of the sample rate,and time in samples. To build a structure that can be integrated intothe trellis form, an estimate of the first symbol time must be providedthat is accurate to one symbol period (±N/2). If the first symbol timecannot be estimated to this accuracy, a separate synchronizationsequence may be required. States are labeled S_(M,i) for the ith statein the Mth stage. In the general case, there may be N starting states,labeled from S_(0,0) to S_(0, N−1). Each state S_(M,t) has 2×Δ_(max)+1(where Δ_(max) represents the maximum number of samples from symbolduration can exceed from nominal symbol duration) transitions leading toconsecutive states starting at S_(M+1,t−Δmax) and ending atS_(M+1,t−Δmax). To further refine the structure, an a-priori estimate ofthe maximum timing error (expressed in samples per symbol) of R can beused. If N is nominal number of samples per symbol, then R=rN where r ispercentage of timing error. The definition of R becomes Δ_(max)=┌R┐Which is an integer. This limits the states at any given trellis stage Mto states starting at S_(M,0−└R×M┘) up to S_(M, N−1+┌R×M┘) where ┌x┘ isthe ceiling of x (the next higher integer), and └x┘ is the floor of x(the next lower integer). This means that at any stage M the stateS_(M,t) corresponds to a symbol that starts at sample time M×N+t. Anexample of this structure for Δ_(max)=1 and N=4 is shown in FIG. 22.

Referring now to FIG. 23, a second, derived structure involves using thesame tree, but windowing it to limit the number of states is shown asthe windowed structure in FIG. 23. For the windowed structure, anadditional parameter W may be defined as the size of the window into thetree. In this way, only W states at any trellis stage M are kept wherethe first state index is defined as B_(M) (the base state). In order tofind B_(M) at time M+1 the window position may be chosen based on theprobability of the states S_(M,B) _(M) through S_(M,B) _(M) _(+Δmax−)1compared to S_(M,B) _(M) _(+W−)1 down to S_(M,B) _(M) _(+W−Δmax).

The result may be called a ‘folded’ structure (also simply the ‘trellis’structure). In this structure there are precisely N states at each stageM. In order to accomplish this, the tree structure may be folded suchthat state S_(M,t) is mapped into state Z_(M,t % N), where % denotes themodulo operator into positive integers (i.e. (−1)% N=N−1). In order tomaintain timing in this structure, each state Z_(M,t) has a mappingvalue t_(m) that the index of the maximum probability state S_(M,k) thatmaps into Z_(M,t). At each stage, t_(m), to transitions from the stateare determined.

In all of these timing structures, this trellis may be combined with thedata trellis and phase trellis to get a combined set of statesS_(M,t,ϕ,D). Each transition out of this state has a triplet of values(Δ_(t),Δ_(ϕ),b) where Δt is the timing change and it takes integervalues between and −Δ_(max) i.e. −Δ_(max),Δ_(max)+1, . . . −1, 0, 1, . .. , Δ_(max)−1, Δ_(max), Δ_(ϕ) is the phase change, and b is the databit. In the case of any binary waveforms (e.g. FM0 and Miller), the datametric for this state and transition is

${Re}\left\{ {e^{- {j{({\varphi + {\Delta \; \varphi}})}}}{\sum\limits_{i = 0}^{N - 1 + {\Delta \; t}}\; {c_{D,b,i} \times {r\left( {{M \times N} + t} \right)}}}} \right\}$

where C_(D,b,i)=d_(D,b,k)×(1−off)+d_(D,b,k+1)×off and d_(D,b,k) is anideal symbol at the nominal sample rate, where

$k = {{\left\lfloor {i \times \frac{N + {\Delta \; t}}{N}} \right\rfloor \mspace{14mu} {and}\mspace{14mu} {off}} = {\left( {i \times \frac{N + {\Delta \; t}}{N}} \right) - {k.}}}$

Note that for the ‘folded’ version, t_(m) is used in place of t.

In order to capture the timing information with symbol timing that candrift over time, another natural structure of the timing state diagramis a tree structure such as shown in 21-2 where the root node isextended. Assume the nominal timing is N samples per symbol. This meansthat the granularity of the timing captured from the tree structure willbe 1/N. In order to improve the performance of the trellis, this can belimited to an arbitrary rate R. To do this, the states that would falloutside the bounds of the expected drift are eliminated. If the statesare numbered for the first stage of the trellis as S₀ to S_(N−1) for thefirst stage, the second stage would be numbered S_(0−Δmax) toS_(N+Δmax), and for stage M they would be S_(0−M×Δmax) to S_(N+M×Δmax).For this Mth stage the states S_(0−M×Δmax) up to but not includingS_(0−┌R×M┐) would be elided and also states above S_(N−1+┌R×M┐). Eachstate in this structure has a T_(s) associated with it that is M×N+i forstate S_(i) at time M.

In order to reduce the complexity of the structure, either a ‘folded’tree or a ‘windowed’ tree can be used. The ‘folded’ tree is a tree wherenode S_(i) is mapped to S_((i+N) % N) and the T_(s) associated with thenode is the T_(s) associated with the maximum state value between themapped states. This means that the transitions become symmetric as in atrue trellis, but the transitions carry both a metric and a time withthem. In a ‘windowed’ tree structure, an arbitrarily sized window ofstates is maintained. This window is selected by comparing theprobabilities of the edge states. In the case of a windowed tree, oneonly needs to keep track of T_(s) for the first state in the window (allother states will be offset linearly from that state). This provides anadvantage of smaller storage and simpler implementation.

Combined Metrics for Data and Timing

$\sum\limits_{i = 1}^{N + {\Delta \; t}}\; {{d_{in}(i)} \times {r\left( {T_{s} + i} \right)}}$

Where

-   -   r(i) is the sample at time i    -   d(i) is the ideal symbol sampled at the sample rate (for the        data state and input data from the trellis)    -   d_(in)(i) is the interpolated version of ideal symbol.    -   T_(s) is the time stored in the state.

$p = {{\left\lfloor {i \times \frac{N + {\Delta \; t}}{N}} \right\rfloor \cdot {off}} = {\left( {i \times \frac{N + {\Delta \; t}}{N}} \right) - p}}$d_(in)(i) = d(p) × (1 − off) + d(p + 1) × off

An example for is given for the case of N=4, R=10%, M=4 in 22-2 and22-4.

Referring now to FIG. 24, a SISO decoder can be viewed as consisting offour consecutive operations:

1) data metric generation and phase rotation

2) Branch metric generation and forward node update

3) Backward pass node update

4) Extrinsic generation and output

Basically, the decoder structure can be viewed as a trellis of nodesarranged in columns. Each column corresponds to one symbol of the datastream to be decoded. The nodes within the columns represent thepossible combinations of the relevant parameters for the symbol; inparticular, the timing, phase, and symbol states. Each contains anumerical value proportional to the computed probability of theparameters which it represents. For FM0, one can use 512 nodes percolumn corresponding to the combinations of the 16 possible (quantized)phase states, the 16 possible timing states, and the two possible symbolvalues (0 and 1). The decoder operates by estimating the probability ofeach node's combination of parameters using metrics derived from theinput data. First, probabilities for the nodes within columns areupdated in the forward time direction, and then in reverse workingbackwards through the trellis. When these updates have been completed,the highest computed probability values through the trellis are chosenfor the decoded output.

The inputs to the SISO decoder trellis computation are the data metricswhich are derived from the sampled input stream. These are complexnumbers, derived from Si sample values containing both I and Qcomponents. Although there are a total of twelve data metrics which mustbe computed for each discrete sample time, N, six of these are simplynegatives of the others as selected by D, the current data symbol state.The metrics, M_(N)(Δt,D,d), where Δt={−1,0,+1} for timing change,D={0,1} for current data state, and d={0,1} for the next symbol value,may be computed from intermediate variables A, B, C, D, E, and F where:

A=Σ S_(i), (n≤i≤n+7)=sum of 8 samples starting at time nB=Σ S_(i), (n+8≤i≤n+15)=sum of 8 samples starting at time n+8C=Σ S_(i), (n≤i≤n+6)=sum of 7 samples starting at time nD=Σ S_(i), (n+8≤i≤n+14)=sum of 7 samples starting at time n+8E=Σ S_(i), (n≤i≤n+8)=sum of 9 samples starting at time nF=Σ S_(i), (n+9≤i≤n+16)=sum of 8 samples starting at time n+9

Referring now specifically to FIG. 24, the Intermediate Metric VariableComputation is shown. The FM0 data metrics M_(N)(Δt,D,d), can then bederived from the intermediate variables as follows:

M_(N)(0,0,0)=A_(N)−B_(N)=−M_(N)(0,1,0)M_(N)(1,0,0)=A_(N)−F_(N)=−M_(N)(1,1,0)M_(N)(−1,0,0)=C_(N)−D_(N)=−M_(N)(−1,1,0)M_(N)(0,0,1)=A_(N)+B_(N)=−M_(N)(0,1,1)M_(N)(1,0,1)=E_(N)±F_(N)=−M_(N)(1,1,1)M_(N)(−1,0,1)=A_(N)+D_(N)=−M_(N)(−1,1,1)

Regarding phase rotation, the node update operation does not use theM_(N) directly, but rather uses the real portion of the complex datametric vector as rotated by the interpolated phase, φ, for each trellisbranch. The rotated data metric is expressed asR_(N)(φ)=Re[M_(N)*e^(jφ)]. Since there are only sixteen evenly spaceddiscrete values for the node phase state, the interpolated branch phasecan only take on 32 values and the product computation is greatlysimplified. This is shown in the table below. Due to symmetry, 16 of thevalues are simply derived by negation of those calculated π radiansaway. This means that all 32 rotations can be computed using 14multipliers and 14 adder/subtractors. By sequencing the 6 values forM_(N) as input into the phase rotator block, all 32 rotations of alltwelve metrics may be computed in real-time at the sample rate. Theoutputs are fed to the node processors for branch metric computationwhere they are added or subtracted as needed.

Regarding backwards pass data metric storage and sequencing, the rotateddata metrics must also be fed to the node update mechanism for thebackwards node update pass. This requires either storage of the valuescomputed for the forward pass, or else regeneration from either the dataor data metrics. In either case storage memory is necessary. Using a256-bit packet for purposes of illustration, storage of the rotated datametrics would require: 16×6×16×256=393,216 (16-bit) words of storage. Atthe other extreme, storage of the interpolated input data stream wouldrequire only 2×16×256=8192 words of storage. While storing theinterpolated data alone would save substantial memory, it requires thatthe metric generation shift-register (as shown above) be run in thereverse direction (from right-to-left), with the stored data fed to itreversed in time, in order to derive the data metrics for the backwardspass. The data resurrected data metric values must the be fed to thephase rotator as before. The R_(N)(φ) outputs must, however, beresequenced for presentation to the node processors. Recall that for thedata metrics, M_(N)(Δt,D,d)=−M_(N)(Δt,˜D,d). This allowed the nodes withD=1 during the forward update pass simply to be fed the negatives of theR_(N)(φ)'s for their D=0 counterparts. During the backwards pass,however, we are indexing the node processors by d instead of D since oursource nodes are now later in time. Consequently, the M_(N)(Δt,D,d) areno longer the arithmetic complements for d=0 vs. d=1; and instead theproper R_(N)(φ) must be stored, sequenced, and fed to the nodeprocessors.

Regarding branch metric generation and node updating, the branchmetrics, B_(XY), where X is the originating node within symbol column C,and Y the destination node in column C+1, are calculated as

B _(XY) =d*S _(C) +R _(N)(φ,Δt,D,d)+U(Δφ),+V(Δt)

Where d=the destination data state (i.e. input data bit value)

S_(C)=Soft input value for column C

R_(N) (φ,Δt,D,d)=Re[M_(N)(Δt,D,d)*e^(jφ)], the rotated data metric

U(Δφ)=one value for Δφ=0, another for Δφ=+1,−1

V(Δt)=one value for Δt=0, another for Δt=+1,−1

For FM0, there are 18 branches out of each source node, corresponding tothe 3 values for Δφ, times the 3 values for Δt, times 2 values for d.Accordingly there are also 18 branches into each destination node. Toupdate the probability score, Q_(Y), for a destination node, the Q_(X)from the source node is added to the branch metric for all inputbranches leading directly to node Y. The value for the branch with thegreatest sum is then selected and stored for Q_(Y). The associatedsample time value, T_(Y), must also be stored, where T_(Y)=T_(X)+16+Δt(or for reverse updates: T_(Y)=T_(X)−16−Δt), and T_(X) is the storedtime value from the source node for the selected branch.

Table for Phase Rotation e^(jφ) Angle e^(jφ) Imaginary Real Part of φReal Part Part Metric Product 0 1 0 x    π/16 d e  dx − by  π/8 a b  ax− by  3π/16 f g   fx − gy  π/4 c c  cx − cy  5π/16 g f  gx − fy 3π/8 b a bx − ay  7π/16 e d  ex − dy  π/2 0 1 −y  9π/16 −e  d −ex − dy 5π/8 −b a −bx − ay  11π/16 −g  f −gx − fy  3π/4 −c  c −cx − cy 13π/16 −f  g  −fx− gy 7π/8 −a  b −ax − by 15π/16 −d  e −dx − gy π −1  0 −x 17π/16 −d  −e −dx + ey  9π/8 −a  −b  −ax + by 19π/16 −f  −g   −fx + gy 5π/4 −c  −c −cx + cy 21π/16 −g  −f  −gx + fy  11π/8  −b  −a  −bx + ay  23π/16 −e −d  −ex + dy 3π/2 0 −1  y 25π/16 e −d   ex + dy 13π/8  b −a   bx + ay27π/16 g −f   gx + fy 7π/4 c −c   cx + cy 29π/16 f −g    fx + gy 15π/8 a −b   ax + by 31π/16 d −e   dx + ey Data Metric M_(N) = x + jy a =cos(π/8), b = sin(π/8) c = sin(π/4) = cos(π/4) d = cos(π/16), e =sin(π/16) f = cos(3π/16), g = sin(3π/16)

Regarding the data-driven update mechanism, the 16 values output fromphase rotation for the rotated data metrics, R_(N)=Re[M_(N)*e_(jφ)] forD=0; φ=0, π/8, π/4, 3π/8, 3π/2, 13π/8, 7π/4, and 15π/8, and specified N,d, and Δt; are fed to the node update mechanism. Since the remainingphase angles can also be derived from these 16 simply by negating thecorresponding value π radians away; and since the values of the R_(N)for D=1 are also just the negatives of those for D=0; each set of theeight R_(N) values is sufficient for metric generation for the threebranches (Δφ=−1, 0, +1) from up to 32 source nodes (all nodes of thespecified time state N) in the originating symbol column, C. These R_(N)values may be labeled as negated for specified values of φ and D asR_(NφD). For forward update, the R_(NφD) are fed into 32 nodeprocessors. These processors compute the branch metrics B_(XY) and sumthem with the stored source node values, Q_(X). The branch sums arepassed to 16 branch selection units which compare six input branchvalues and select the largest for output. Each selection unitcorresponds to a specific phase value, φ_(Y). The inputs are then thebranches where φ_(Y)=φ_(X)−(Δφ*π/8), including the branches for both d=0and d=1. The outputs of these selection units then feed back to the twonode processors of corresponding phase where they are used to update thestored Q_(Y) value for the destination node. A minimum of ninety-sixclocks are required to update each symbol column. For the reverse updatedirection, the rotated metrics are regenerated starting with the mostrecent symbol, and proceeding back to the first. The later column nodes(to the right in the trellis) are used for the source values, Q_(X), andthe earlier column to the left are now updated as the Q_(Y)

Referring now to FIG. 25, the interconnection of node processors andbranch select units are shown. For convenience, upwards arrows to Δφ=+1for forward updates, −1 for backwards updates. Downwards arrows are forbranches with Δφ=−1/+1 for forward/backward updates. Horizontal arrowsare for Δφ=0.

Referring now to a source node processor, the proposed implementationconsists of 32 node processors, each assigned to a particular data statevalue (0 or 1), and phase state (0 to 15). One of the eight R_(N)values, or its arithmetic complement, as appropriate, are fed into eachnode processor corresponding to its assigned data and phase statevalues. Each processor consists of a node memory, a comparator, and fouradders. This structure is shown in the diagram below. The node memorystores two values for each node, the probability score Q, and the timecode T. Each processor's node memory contains all trellis nodes for aspecific data state, D, and for a specific phase state value, φ. It alsocontains the storage for the nodes in all trellis symbol columns, C, andfor all sixteen time state values, N. For a 256 symbol decoder,16×256=4096 node storage locations would be required. Local storage canbe greatly reduced for large message sizes if paging to external memoryis implemented.

The adders function to generate three branch metrics on each clock asthe six R_(N)'s are fed in sequentially for each N. Therefore, a minimumof 96 clock cycles are required to update a symbol column. The addersserve to sum the various terms for the branch metric values B_(XY) withthe source node value, Q_(X). The value d*S_(C), being global to allprocessors, is developed externally. The value for V(Δt) is alsoselected outside and summed with d*S_(C) to be fed to the nodeprocessors as an input. Remaining inputs are the symbol column number C,which is concatenated with the timing sample state N to address aparticular node within local storage; the rotated metric value R_(NφD),and the two values for U(Δφ).

Referring now to destination node processing, as the R_(NφD)(d,Δt) arefed into the source node processors, sequentially stepping through thesix combinations of d and Δt for each sample time N, the B_(XY)+Q_(X)sums are output to the destination nodes for comparison and selection.Since d is fixed at each clock, there are only sixteen destination nodesfor the 96 branches generated on each clock. This means there are sixpotential branches into each destination node at each clock which needto be compared and selected for the maximum. Along with the branch sum,the corresponding time value, T_(X), from the source node for thewinning branch must also be selected and stored. Destination processingcan be performed by a seven-input maximum selector. The seventh input isused to compare any previous maximum partial results from the threeupdate cycles required to examine all eighteen of the branches into adestination node. The results of each of these sixteen selectors isshared as input to the two node memories sharing the same time statevalue, N, but one with D=0, and one with D=1. It should be noted thatthe destination node N_(Y) time-state value is not necessarily the sameas the source N_(X) value, but is rather equal to (N_(X)+Δt) modulo 16.

Referring now to time divergence, a possible problem with the basicsource node processing as shown in the diagram above lies in the way inwhich the trellis tracks timing. There are several timing variables ofinterest. T refers to absolute sample time as numbered sequentially fromthe first data sample. Each node in the trellis also has a fixed 4-bittiming state value, N, ranging from 0 to 15. This 4-bit value alwayscorresponds to the LS 4-bits of the absolute sample time assigned tothat particular node. That assigned T can, however, change within thesymbol column depending upon T_(XY) value for the branch selected duringnode update, where T_(XY)=T_(X)+Δt. This T_(XY) value should thereforebe stored in the node when it is updated. When generating the branchmetric values, it may be necessary to compare the stored T_(X) for thenode with the sample time T_(NφD), as it is possible for the stored Tassigned to a node with timing state N, to be different. This means thatwith the basic architecture of the diagram, multiple passes may beneeded to present the rotated data metric for all node T values at agiven N, φ, and D in order to generate all of the branch metrics. Thisis the reason for the equality comparator and the valid line shown inthe diagram. In order to increase parallelism and reduce the number ofclock passes required, it is highly desirable to present severalpossible rotated metrics in parallel to the node processor so thatbranches for varying T's, but with specified N, can be generatedsimultaneously. Source node processing architecture for multipleparallel T updates is shown below. Since total divergence is limited bythe length of the data packet, and since the LS-4 bits are redundant, itis not necessary that all bits of T_(XY) be stored and compared in thenode processor. The additional R_(NφD) can be made available by savingthe 8 rotated metric values generated for each clock in delay storage oflength 96 clocks for each additional R_(NφD)(T) to be presented. Thisrequires four 18K block RAM for every two additional values of T, if theRAM is operated at the same clock rate as the branch metric generator.

Referring now to FIG. 26, the extended parallel source node processingis shown.

Referring now to FIG. 27, the forward and backward processing is shown.

Data Phase φ State State Forward R_(Nd)(φ) Backward R_(Nd)(φ) 0 0 0R_(NX)(0)  R_(N0)(0)  0 1 0 −R_(NX)(0)    R_(N1)(0)   π/8 0 1R_(NX)(π/8) R_(N0)(π/8)  π/8 1 1 −R_(NX)(π/8)  R_(N1)(π/8)  π/4 0 2R_(NX)(π/4) R_(N0)(π/4)  π/4 1 2 −R_(NX)(π/4)  R_(N1)(π/4) 3π/8 0 3 R_(NX)(3π/8)  R_(N0)(3π/8) 3π/8 1 3 −R_(NX)(3π/8)  R_(N1)(3π/8)  π/2 04 −R_(NX)(3π/2) −R_(N0)(3π/2)  π/2 1 4  R_(NX)(3π/2) −R_(N1)(3π/2) 5π/80 5  −R_(NX)(13π/8)  −R_(N0)(13π/8) 5π/8 1 5  R_(NX)(13π/8) −R_(N1)(13π/8) 3π/4 0 6 −R_(NX)(7π/4) −R_(N0)(7π/4) 3π/4 1 6 R_(NX)(7π/4) −R_(N1)(7π/4) 7π/8 0 7  −R_(NX)(15π/8)  −R_(N0)(15π/8)7π/8 1 7  R_(NX)(15π/8)  −R_(N1)(15π/8) π 0 8 −R_(NX)(0)   −R_(N0)(0)    π 1 8 R_(NX)(0)  −R_(N1)(0)    9π/8 0 9 −R_(NX)(π/8) −R_(N0)(π/8)  9π/8 1 9 R_(NX)(π/8) −R_(N1)(π/8)  5π/4 0 10 −R_(NX)(π/4) −R_(N0)(π/4)  5π/4 1 10 R_(NX)(π/4) −R_(N1)(π/4)  11π/8  0 11−R_(NX)(3π/8) −R_(N0)(3π/8) 11π/8  1 11  R_(NX)(3π/8) −R_(N1)(3π/8) 3π/20 12  R_(NX)(3π/2)  R_(N0)(3π/2) 3π/2 1 12 −R_(NX)(3π/2)  R_(N1)(3π/2)13π/8  0 13  R_(NX)(13π/8)  R_(N0)(13π/8) 13π/8  1 13  −R_(NX)(13π/8) R_(N1)(13π/8) 7π/4 0 14  R_(NX)(7π/4)  R_(N0)(7π/4) 7π/4 1 14−R_(NX)(7π/4)  R_(N1)(7π/4) 15π/8  0 15  R_(NX)(15π/8)  R_(N0)(15π/8)15π/8  1 15  −R_(NX)(15π/8)  R_(N1)(15π/8)

Extrinsic generation may be performed as the nodes are updated in thereverse direction. A reliability measure is also computed. The extrinsicis computed as Max(α_(X)+B_(XY)+β_(Y)) over each column.

It is clear to a person having ordinary skill in this art that thetechniques described above may be applied to a communication method orsystem for processing a modulated signal with random data, and/or phaseand/or unknown timing to estimate received data sequences or packetizeddata. The receiver may use iterative processing withsoft-input-soft-output (SISO) components to combine channel decodingwith equalization, demodulation, phase tracking, symbol timing,synchronization and interference cancellation as part or in whole. Thesetechniques may be used for any wireless communication systems to modelthe observation space. These techniques may be used for sensory receiversystem for detecting signals in presence of noise and channel distortionutilizing iterative method to detect the signal. A communication systemmay use these techniques for maximum likelihood sequence estimationwhich may include lattice, trellis or tree structures or productsthereof for joint estimation of phase, timing, data and/or baud rate.Such techniques may also be used in signal detection systems utilizingiterative methods for optimal detection of the signal in the presence ofwhite noise and channel distortion such as those employing in-doorwireless channels, out-door wireless channels, both line-of-sight ornon-line of sight communications, wire line channel such as copper andfiber wires, underground or underwater sonar, recording channels such ashard disk storage and both volatile and non-volatile memory and/orcombinations of any of these channels.

The disclosed techniques are useful in the detection of packetized datawith unknown data pulse duration, random phase and unknown data, anycombination or thereof. They are useful in digital packet radio systemsemploying soft-input-soft-output (SISO) decoding methods with or withoutcascaded iterative decoder and with or without channelencoding/decoding. These techniques may be used in communication systemsemploying channel coding methods including algebraic block codes,convolution and turbo codes, low density parity check, repeat-accumulatecodes, and product codes cascaded with SISO decoders exchangingextrinsic information to optimally decode the user data. Similarly,these techniques may be used in communication systems employing channelcoding including coding which can be represented via planar graph suchas bipartite, tree or trellis diagram whereby the posterioriprobabilities (extrinsic information) of each state can be computed anditeratively improved. Such communication systems may employing beliefpropagation method to decode the received sequence and exchangeextrinsic information with the soft-input-soft-output decoder. Acommunication system or packet radio timing synchronization may beprovided for any modulation scheme such as multi-level phase, position,amplitude in quadrature and in-phase (one or both). Further, suchsystems may be embedded in portable or stationary devices, in hubs,central office or network edge devices and may be implemented insoftware, such as a “Software Defined Radio”, on special purpose orgeneral purpose host computer and offered as a web service or generalpurpose signal processing platform.

Referring now to FIG. 28, in a preferred embodiment, a digital transmitwaveform generator 28-05 suitable for low complexity implementation maybe used to generate an arbitrary symbol waveform 28-07 that is symmetricin time, for example, for use as a waveform generator for RF modulationsemployed by the EPC Global Specification for RFID Air Interface. Forthis RFID application, the waveforms may be single-sidebandamplitude-shift keying (SSB-ASK), double-sideband amplitude-shift keying(DSB-ASK), and Phase-reversal amplitude-shift keying (PR-ASK).

Referring now to FIG. 28, a high-level block diagram of waveformgenerator 28-05 is shown. The type of waveform to be generated may beselected by operation of controller 28-10. For RFID applications, theselectable waveforms may be SSB-ASK, DSB-ASK, and PR-ASK. Pulses of6.25, 12.5 and 25 microseconds are accommodated. The waveform generator28-05 may be frequency agile so that it can be used with frequencyhopping and or FDMA mode of operations. The frequency range andresolution is a function of the logic clock as well as the bitprecisions used in the implementation of waveform generator 28-05. Anarbitrary phase offset can also be added to the waveform pulse.Frequency and phase may be selected via controller 28-10.

The reduced complexity waveform generation architecture may include aWaveform Look Up Table (LUT) 28-20 and a Ramp-Up/Ramp-Down block 28-30.Portions of waveforms, such as waveform samples 28-40, may be stored inLUT table 28-20. The LUT table 28-20 may store only half of thesteady-state portion, i.e. LUT portion 28-50, of sample waveform 28-40.The other half of the waveform, mirror image portion 28-60, may begenerated by up-down block 28-30 and therefore need not be stored.

In operation, in response to a control signal from controller 28-10,waveform LUT 28-20 applies the LUT content, such waveform sample 28-50,to ramp-up/down block 28-30 which generates mirror image 28-60, thetransient portion of the waveform 28-70, in hardware. When compared to aconventional waveform generator, the memory usage of waveform look uptable 28-20 is drastically reduced. This reduction comes from two areas.First, the number of waveform sample points is reduced by more than afactor of 2 because only one half of the waveform sample need be stored.To be precise, the storage may be (½−T_transient/Tsym) of the fullwaveform, where T_transient is the transient duration and Tsym is thesymbol time. More importantly, the sampling spacing of the waveform inLUT 28-20 can be reduced to the transient duration. This is because theramp-up/ramp-down block 28-30 generates the transient portion (mirrorimage 28-60) of waveform 28-70 using the higher sampling rate of thewaveform generator output.

In RFID applications, the quadrature (Q) channel of the waveform symbolto be generated may be a constant. Hence, ramp-up/ramp-down block 28 maybe required for the in phase (I) channel only.

A conventional numerically controlled oscillator (NCO) 28-80 may be usedto generate the in-phase and quadrature IF carriers, with the desiredfrequency and phase offset, in response to controller 28-10. The primarypurpose of NCO 28-80 is to shift the IF signal frequency within thedesired passband so that the SSB signal can be centered on the nominalcarrier frequency, without having to shift the actual fixed frequencyplan of the RF up-conversion. The quadrature carriers are fed into thein-phase and quadrature (I-Q) mixer 28-90 to generate the IF waveform,I′ and Q′, with the appropriate symbol. These signals may be applied toa transmitter ADC.

In many embodiments, the symbol rate of the symbols transmitted by anRFID tag can vary. By constraining the symbols that can be transmittedby an RFID tag to a predetermined standard, knowledge of the imposedconstraints can be used to estimate the symbol rate of the informationtransmitted by the RFID tag. Referring now to FIG. 29, an illustrationof synchronizing bursty data, with application to an RFID system, isshown. In the RFID standard, received signal 29-10 may consist of apilot tone 29-12, preamble 29-14, and data 29-16. The pilot tone 291-12may consist of 12 “zero” symbols and preamble 29-14 may be a fixedpattern of 6 symbols. Data 29-16 may include “n” data symbols to beprocessed by the data burst synchronizer. Note that n can encompass thewhole data sequence but it can also be a subset of the sequence.

Since for the RFID standard FM0 and Miller codes, each symbol may onlyswitch sign in the middle, it is convenient to consider half-symbols,such as half symbols h23 and h22, representing the half symbols or(binary h_(k)'s) of the first “0” in pilot tone 29-12 in FIG. 29. Forthe FM0 code, every symbol transition also introduces a sign change.

The task of the data burst synchronizer is to reconstruct the timing,such as reconstructed timing 29-20, from received signal 29-10, tofacilitate data detection and processing. This task includes finding thestart time of the data burst (identified by the end of preamble 29-14)and matching the reconstructed symbol clock frequency with the receivedsymbol clock frequency, or baud rate, of received signal 29-10. If thereconstructed timing 29-20 is not aligned with the received signal, itmay have a timing error 29-30 which may consist of s integer and T□fractional half-symbols. Here T is normalized to the half-symbol timeT_(s)/2. Error in the reconstructed baud rate may result in insertion ofextraneous, or deletion of desired, data symbols as indicated by thecumulative effect of imperfect baud rate 29-40 illustrated at the end ofdata 29-16 shown in reconstructed timing 29-20 in the figure.

An asynchronous data burst synchronizer, suitable for digitalimplementation, may be optimized by maximizing a metric that is afunction of the reconstructed sample baud rate and timing error. Themetric may be generated by correlating a known data structure with thereceived data burst signal 29-10, which may include pilot tone 29-12,preamble 29-14, and data 29-16. The optimum received signal timing andclock frequency can be found by correlating a replica of the knownportion of the transmit data waveform with a set of timing and clockfrequency hypotheses spanning the frequency and timing uncertainty ofthe arriving signal. The best hypothesis will yield the highestcorrelation metric. Since every symbol transition introduces a signchange in FM0, the optimum correlator also takes advantage of thisinformation. The following metric may be used for the FM0 code:

$\begin{matrix}{{M\left( {b,\tau,s} \right)} = {{{{\sum\limits_{k = {- 23}}^{0}\; {{r_{k + s}\left( {b,\tau} \right)}h_{k}}} + {\sum\limits_{k = 1}^{12}\; {{r_{k + s}\left( {b,\tau} \right)}h_{k}}}}} + {\sum\limits_{k = 7}^{n + 6}\; {{{r_{{2k} + 1 + s}\left( {b,\tau} \right)} - {r_{{2k} + s}\left( {b,\tau} \right)}}}}}} & (42)\end{matrix}$

where r_(i)(b, i) denotes the reconstructed half-symbol at time i usinga baud rate b and the timing hypothesis s+τ, h_(k) is the binaryhalf-symbols for the pilot and preamble. The first term in equation (42)is the pilot correlation for the 2×12 half-symbols and the preamblecorrelation for 2×6 half-symbols. The second term is a correlation ofthe sign change during the symbol transitions of the FM0 code. This termis not used for the Miller code. Because of the pilot preamble structureas well as the transition operation, the metric (42) is independent ofany dc bias.

Referring now to FIG. 30, a functional architecture of optimal burstsynchronizer 30-05 is shown, based on maximizing the metric in equation(42). The synchronizer first interpolates I and Q samples 30-10 from thereceiver analog-to-digital converter (ADC) in interpolator 30-20. Aftercorrelating with the fixed pattern including the preamble, an estimateis derived for the baud rate and starting position of the tag responsedata.

The frequency tolerance specified by the EPC spec for the highest-speedstream (640 kbps) is +/−15%. Interpolator 30-20 develops a sufficientnumber of streams of signal data samples at baud rates spaced to spanthe frequency uncertainty range. The interpolator 30-20 performs alinear interpolation of two adjacent input data samples lying closest tothe desired sample point in time from the selected baud rate.

I and Q values of each interpolated streams are correlated with thefixed data pattern in stream storage and pattern correlator 30-30. Thiscorrelation is performed at the selected sample rate for eachinterpolated stream. Since the fixed data pattern is +1/−1, thecorrelation can be accomplished by summing the number of interpolatedsamples for each half-symbol, and then adding or subtracting these sumsdepending upon the expected preamble waveform at the correspondingposition.

The maximum of the metric of all correlator output is chosen by thescorer 30-40 as the best estimate for preamble position in time and alsoas the best estimate of the incoming baud rate of the tag response. Theinterpolated I-Q sample values for the chosen sample rate are then usedto compute the data-metrics for SISO processing. These are then fed to aSISO block, as shown above, for example, in FIGS. 2b , 3, 4, 9 c, 9 d,10, 11 and 12 b.

Referring now to FIG. 31, an interpolation algorithm suitable fordigital implementation, using a digital differential delay analyzer andtwo multipliers, for appropriately weighting the nearest input samplevalues is shown. Interpolator 30-20 generates re-clocked samples 31-20at the desired rate 1/T_(i) from the incoming received signal samples31-10 which are at a fixed rate of 1/T. In this illustration, thereconstructed signal is at a lower sampling rate. An interpolated sampleis a weighted sum of the two nearest neighbor of the incoming signalstream. For example, the interpolated signal sample y₂ is computed viathe weighted sum in computer block 31-30. In general, the interpolationcan be accomplished with the recursions:

y _(m+1) =F _(m+1) x _(p)+(1−F _(m+1))x _(p+1)

F _(m+1)=frac(S _(m+1))

p=└S _(m+1)┘

S _(m+1) =S _(m) +T _(i) /T

S ₁=0;F ₁=0;m=1,2, . . .  (43)

where F_(m) is the weighting factor for the closest left incoming sampleand (1−F_(m)) is the weighting factor for the right sample. In equation(43), frac (•) denotes the fractional part of a real value and └•┘denotes the integer part of a real value. In the digital differentialdelay analyzer, the ratio T_(i)/T is added to an accumulator S everyT_(i). Then F_(m) and p are computed from the accumulated sum asindicated in equation (43) above. The process is similar if thereconstructed symbol clock frequency is higher than the input, exceptmore than a single interpolated sample may need to be generated duringan input sample period.

Referring now to FIG. 32, a more detailed block diagram of thecorrelator block 30-30 in FIG. 30 is shown. To reduce the number ofcomputations, the interpolated stream is first summed over a half-symbolworth of signal samples in adder 32-10. The half-symbol sum is thencorrelated with the pilot pattern 32-20, the preamble pattern 32-30, andfor FM0 only, the symbol transition pattern 32-40. The results areproperly delay-matched, for example, in 6+n symbol delay 32-22 and nsymbol delay 32-32 and summed in adder 32-24 and combined with theoutput of transition accumulator 32-40 in adder 32-44 to form thedesired metric 32-50. Note that since the computation is at theinterpolated sample rate, half-symbol correlation corresponding to adifferent sample start time may be generated every sample clock. Desiredmetric 32-50 may be stored in memory location 32-60 where s is the starttime for half symbols and T is the start time offset in fractions of ahalf symbol.

Referring now to FIG. 33, two digital building blocks are shown thatcould be used to implement the functional blocks illustrated in FIG. 31.The half-symbol summation can be implemented with the difference and sumblock 33-10. The delay (in number of signal samples) may be selected tomatch the half-symbol time. Since the fixed half-symbol pattern is +/−1,the pilot and preamble correlator can be implemented with delayregisters and accumulators as shown in correlator 33-20. Here the shiftregister positions corresponding to +1 fixed pattern are summedseparately than those corresponding to −1. Then the −1 intermediate sumis subtracted from the +1 intermediate sum in the last step. Thetransition correlator can be implemented using the basic structure ofdifference and sum block 33-10 in which case the delay is set to match asymbol time.

Referring now to FIG. 34, an illustration of a pallet code techniqueused to read RFID tags blocked by obstructions is shown. FIG. 34illustrates tag blockage in an RFID system such as reading a batch oftags affixed to merchandise sitting on a pallet. The RFID tags on pallet34-10 are to be scanned by a reader 34-20 but some of the tags areblocked from its view by an obstruction 34-30. The goal is toreconstruct lost data in the blocked tags by reading the remainingnon-blocked tags.

A “Pallet code” for reading blocked RFID tags is disclosed herein.Information for each tag in a batch may be shared among all tags in thebatch to be read in the form of redundancy provided by the codingscheme. The redundant bits could be stored in the reader and used laterto read the tags. Alternatively, the redundant bits could be distributedamong the tags by appending the bits to the tag data packet.

Referring now to FIG. 35, one implementation of a Pallet code isdisclosed. Assume the batch of tags to be read, for example a group oftags on units contained on pallet 34-10, consists of M tags and each tagto be read contains a data packet Pi of n bits as shown in data packetgroup 35-40 including data packets P₁ to P_(M). The same encoder 35-10receives M information bits 35-20 from bit position j of each datapacket in data packet group 35-40 and produces L redundant bits 34-60.The resulting M•L redundancy bits could be stored in the reader 34-20 orequally shared by the data packets in data packet group 35-40 byappending the additional redundant data to the tag data packet fromgroup 35-40. The decoder in RFID tag reader 34-20 uses the redundantbits to reconstruct data from the tags blocked by obstruction 34-30shown in FIG. 34.

To conserve bandwidth, it is desirable to limit the L overhead bits to asmall number so that the code rate R=M/(M+L) is close to 1, say between0.7 and 0.99. Yet the code correction capability improves withredundancy. Theoretically, for large M we should be able to correct allblocked packets if the probability of blockage is independent and lessthan (1−R). The selection of the code rate R is a tradeoff betweenavailable bandwidth and expected blockage environment.

If the number of tags M is small, it may be preferable to use shortblock-length high-rate codes such as BCH or RS codes because of lowerdecoder complexity. If M is more than 500, high rate LDPC codes may bemore attractive. For example, if M=999 and L=111 for a code rate of0.889, a (4, 36) regular LDPC code could be used.

Referring now to FIG. 36, a simulated performance of this code, using asimple message-passing algorithm, is shown on bipartite graph 36-10showing the probabilities of error, as a function of the probability ofblockage, for single data packets 36-20 and all data packets 36-30. Forexample, it shows that all tags can be recovered with 10⁻⁶ probabilityof error if the probability of blockage is 4%. In the simulation, abinary symmetric channel is assumed—the variable nodes in the bipartitegraph are either correct (not-blocked) or erased (blocked).

Referring now to FIG. 37, the simple message-passing decoding algorithmused in the simulation is shown. In summary, for each such check,correct the corresponding erased node by adding all bits of correctnodes connected to that check using an Exclusive OR operation. If thisset is empty, declare failure and end the algorithm. In particular, instep 37-10, the checks and corresponding edges are removed if all edgesfrom these checks are connected to the correct nodes. In step 37-20, ifall checks in bipartite graphs are removed, declare correction and endthe algorithm. In step 37-30, consider set of checks with only oneerased node connected to the check. In step 37-10, the algorithm returnsto step 37-10 to process any remaining checks. This algorithm isamenable to low complexity implementation.

1.-20. (canceled)
 21. A receiver configured to decode a data signalmodulated onto a carrier wave, where the phase and timing of the datasignal is ambiguous, comprising: an analyzer and equalizer configured tofilter an input signal; an estimation block configured to obtain abaseband representation of the modulated data signal by mixing thefiltered input signal with the carrier wave; and a detector configuredto perform timing recovery on the modulated data signal in the presenceof noise and to determine a sequence of data symbols; wherein: thedetector includes an interpolator that is configured to sample andinterpolate the modulated data signal to generate a plurality of streamsof signal samples possessing different symbol rates; the detectorincludes a correlator that is configured to: for each interpolatedstream of signal samples: sum the interpolated stream over a half-symbolworth of signal samples; correlate the half-symbol sum with a pilotpattern; correlate the half-symbol sum with a preamble pattern; anddelay match and sum the pilot pattern correlated result and the preamblepattern correlated result; and select the stream having the maximumcorrelation.
 22. The receiver of claim 21, wherein the analyzer andequalizer is configured to filter at least one source of narrowbandinterference from the input signal.
 23. The receiver of claim 22,wherein the analyzer and equalizer includes a low latency notch filter,where the location of the notch can be moved to eliminate sources ofnarrowband interference from the input signal.
 24. The receiver of claim23, wherein: the notch filter is implemented using a filter bank with animpulse response determined by a set of filter bank coefficients; andthe analyzer and equalizer estimates the channel impulse response anduses it to determine the filter bank coefficients.
 25. The receiver ofclaim 23, wherein the notch filter is configured to adapt the locationof the notch based upon an output of the detector.
 26. The receiver ofclaim 21, wherein the estimation block receives the carrier wave as aninput.
 27. The receiver of claim 21, wherein the estimation block isconfigured to estimate the frequency of the carrier wave.
 28. Thereceiver of claim 27, wherein: the estimation block is configured tocontrol a programmable oscillator; and the estimation block isconfigured to estimate the frequency difference between the transmittedcarrier wave and the output of the programmable oscillator and toreconfigure the programmable oscillator to reduce the frequencydifference.
 29. The receiver of claim 21, wherein the detector includesa decoder that determines the sequence of symbols with the maximum aposteriori probability of having been transmitted given the data signal.30. The receiver of claim 29, wherein the decoder is configured using afinite state machine to model the observation space.
 31. The receiver ofclaim 30, wherein the finite state machine incorporates symbol phaseestimation.
 32. The receiver of claim 30, wherein the finite statemachine incorporates symbol timing estimation.
 33. The receiver of claim29, wherein: the data signal is channel coded; the decoder comprises: asoft metric estimator; a de-interleaver; a soft input soft output (SISO)decoder; an interleaver; and a channel code decoder; the soft metricestimator is configured to calculate initial soft metrics using the datasignal and a fixed phase value, timing value and channel state estimatedby the channel code decoder during a previous iteration; thede-interleaver is configured to de-interleave an input generated bysubtracting the output generated by the interleaver in a previousiteration from the initial soft metrics; the SISO decoder is configuredto generate updated soft metrics using the output of the de-interleaver;the interleaver is configured to interleave an input generated bysubtracting the output of the de-interleaver from the updated softmetrics; the channel code decoder is configured to estimate a phasevalue, a timing value and channel state from the output of theinterleaver; and the decoder is configured to iterate until the initialsoft metrics and the updated soft metrics converge.
 34. The receiver ofclaim 33, wherein: the decoder determines the maximum soft metric andoutputs the maximum soft metric; and decoder uses predeterminedprobabilities to augment at least some of the maximum soft metrics. 35.The receiver of claim 33, wherein the channel code decoder includes asoft input soft output forward error correction decoder.
 36. Thereceiver of claim 21, wherein: the sequence of symbols includes apreamble known by the receiver; the detector includes an interpolatorthat is configured to sample and interpolate the data signal to generatea plurality of streams possessing different symbol rates; the detectorincludes a correlator that is configured to select a stream using atleast the correlation between the stream and the known preamble; and thedetector is configured to provide the selected stream to a decoder. 37.The receiver of claim 36, wherein: at least a portion of the sequence ofsymbols is constrained to a predetermined set of allowed symboltransitions; and the correlator is configured to select a stream usingat least the correlation between the stream and the known preamble andthe correlation between the stream symbol transitions and the allowedsymbol transitions.
 38. A system for interrogating radio frequencyidentification (RFID) tags, comprising: an exciter in a first locationconfigured to activate an RFID tag; and a receiver in a second locationfor receiving information from an activated RFID tag; wherein theexciter and the receiver are configured to communicate via at least onewireless link; and wherein the receiver is configured to provideinformation to transmit to an activated RFID tag that is responsive toinformation decoded from signals received from the activated RFID tag tothe exciter via the wireless link.
 39. The system of claim 38, whereinthe receiver is configured to perform phase and timing recovery on asignal received from an activated RFID tag.
 30. The system of claim 38,wherein the receiver is configured to decode information received froman activated RFID tag by determining the sequence of symbols with themaximum a posteriori probability of having been transmitted based uponthe signal received from the activated RFID tag.
 31. The receiver ofclaim 21, wherein the correlator is further configured to sum the outputof a transition accumulator with the pilot pattern correlated result andthe preamble pattern correlated result.
 32. The receiver of claim 21,wherein the correlator is further configured to store metrics associatedwith the selected stream using a computed start time for half symbolsand a computed start time offset in fractions of a half symbol.